Apparatus for transmitting broadcast signals, apparatus for receiving broadcast signals, method for transmitting broadcast signals and method for receiving broadcast signals

ABSTRACT

An apparatus for transmitting broadcast signals, an apparatus for receiving broadcast signals, a method for transmitting broadcast signals, and a method for receiving broadcast signals are disclosed. A method for receiving a broadcast signal comprising receiving a broadcast signal, parsing a second packet at the received broadcast signal, parsing the first packet using the fixed header and/or the extended header and displaying broadcast contents by processing the parsed first packet is disclosed.

TECHNICAL FIELD

The present invention relates to an apparatus for transmitting broadcastsignals, an apparatus for receiving broadcast signals and methods fortransmitting and receiving broadcast signals.

BACKGROUND ART

Transmission/reception (Tx/Rx) of IP-based broadcast signals has beenextended in digital broadcast systems. Specifically, the importance ofIP-based broadcast signal Tx/Rx environments has been emphasized in themobile digital broadcast schemes, for example, DVB-NGH from amongEuropean broadcast standards or ATSC-MH from among North Americanstandards. In addition, it is expected that a hybrid broadcast systemdesigned to be interoperable with the broadcast network or the Internetnetwork will be constructed in the next generation broadcast system.

Meanwhile, it will take a long time to fully switch from the legacyMPEG-2 TS based broadcast system to the IP broadcast system in terms ofindustrial or political aspects, and there is a need to develop a newbroadcast system capable of simultaneously supporting IP and MPEG-2 TSschemes.

DISCLOSURE Technical Problem

An object of the present invention is to provide a structure of a linklayer packet that is capable of being processed irrespective of packettypes received from a higher layer in the next generation broadcastsystem.

Another object of the present invention is to provide a method forallowing information contained in packets of a higher layer in aprotocol stack to be used in the packet processing of a lower layer.

Another object of the present invention is to provide a method forreducing overhead during transmission of higher layer packets.

Technical Solution

The object of the present invention can be achieved by providing amethod for receiving a broadcast signal comprising: receiving abroadcast signal, parsing a second packet at the received broadcastsignal, wherein the second packet includes a fixed header, an extendedheader having a variable length, and a payload including the firstpacket, wherein the fixed header includes a packet type element foridentifying a category of the first packet and an indicator part elementfor identifying a construction of the payload, wherein the indicatorpart element includes a C/S (Concatenation/Segmentation) field foridentifying a format of the first packet contained in the payload of thesecond packet, parsing the first packet using the fixed header and/orthe extended header and displaying broadcast contents by processing theparsed first packet.

Preferably, the packet type element identifies that the first packet isan IPv4 (Internet Protocol version 4) packet.

Preferably, the parsing of the second packet at the received broadcastsignal includes: if the C/S field indicates that a general IPv4 packetis contained in the payload of the second packet, parsing a length fieldcontained in a header of the first packet that is spaced apart from apayload start point of the second packet by a predetermined offset andidentifying a total length of the second packet by adding a headerlength of the second packet to a length of the first packet indicated bythe length field.

Preferably, if the C/S field indicates that a plurality of concatenatedIPv4 packets are contained in the payload of the second packet, theindicator part element further includes a count field indicating thenumber of first packets contained in the payload.

Preferably, the parsing of the second packet at the received broadcastsignal includes: if the C/S field indicates that a plurality ofconcatenated IPv4 packets are contained in the payload of the secondpacket, parsing a length field contained in a header of the first packetthat is spaced apart from a payload start point of the second packet bya predetermined offset, and parsing a length field contained in a headerof a next first packet that is spaced apart from a payload end point ofthe first packet identified by the parsed length field by thepredetermined offset, repeating the parsing step of the length fielduntil all the length fields contained in the headers of as many firstpackets as the number of first packets identified by the count field arecompletely parsed and identifying a total length of the second packet byadding a header length of the second packet to the lengths of the IPv4packets indicated by the parsed length fields.

Preferably, the packet type element identifies that the first packet isan IPv6 (Internet Protocol version 6) packet.

Preferably, the parsing of the second packet at the received broadcastsignal includes: if the C/S field indicates that a general IPv6 packetis contained in the payload of the second packet, parsing a length fieldcontained in a header of the first packet that is spaced apart from apayload start point of the second packet by a predetermined offset, andidentifying a total length of the second packet by adding a sum of aheader length of the first packet and a header length of the secondpacket to a length of the first packet indicated by the length field.

Preferably, if the C/S field indicates that a plurality of concatenatedIPv6 packets are contained in the payload of the second packet, theindicator part element further includes a count field indicating thenumber of first packets contained in the payload.

Preferably, the parsing of the second packet at the received broadcastsignal includes: if the C/S field indicates that a plurality ofconcatenated IPv6 packets are contained in the payload of the secondpacket, parsing a length field contained in a header of the first packetthat is spaced apart from a payload start point of the second packet bya predetermined offset, and parsing a length field contained in a headerof a next first packet that is spaced apart from a payload end point ofthe first packet identified by the parsed length field by thepredetermined offset, repeating the parsing step of the length fielduntil all the length fields contained in the headers of as many firstpackets as the number of first packets identified by the count field arecompletely parsed and identifying a total length of the second packet byadding not only a specific value obtained when a header length of theIPv6 packet is multiplied by the number of packets identified by thecount field but also a header length of the second packet to the lengthsof the IPv6 packets indicated by the parsed length fields.

In accordance with another embodiment of the present invention, anapparatus for receiving a broadcast signal comprises: a receiverconfigured to receive a broadcast signal, a common protocol stackprocessor configured to parse a second packet at the received broadcastsignal, wherein the second packet includes a fixed header, an extendedheader having a variable length, and a payload including the firstpacket, wherein the fixed header includes a packet type element foridentifying a category of the first packet and an indicator part elementfor identifying a construction of the payload, wherein the indicatorpart element includes a C/S (Concatenation/Segmentation) field foridentifying a format of the first packet contained in the payload of thesecond packet, and configured to parse the first packet using the fixedheader and/or the extended header, and an A/V (Audio/Video) processorconfigured to display broadcast contents by processing the parsed firstpacket.

Preferably, the packet type element identifies that the first packet isan IPv4 (Internet Protocol version 4) packet.

Preferably, the common protocol stack processor, if the C/S fieldindicates that a general IPv4 packet is contained in the payload of thesecond packet, parses a length field contained in a header of the firstpacket that is spaced apart from a payload start point of the secondpacket by a predetermined offset; and identifies a total length of thesecond packet by adding a header length of the second packet to a lengthof the first packet indicated by the length field.

Preferably, if the C/S field indicates that a plurality of concatenatedIPv4 packets are contained in the payload of the second packet, theindicator part element further includes a count field indicating thenumber of first packets contained in the payload.

Preferably, the common protocol stack processor, if the C/S fieldindicates that a plurality of concatenated IPv4 packets are contained inthe payload of the second packet, parses a length field contained in aheader of the first packet that is spaced apart from a payload startpoint of the second packet by a predetermined offset, and parses alength field contained in a header of a next first packet that is spacedapart from a payload end point of the first packet identified by theparsed length field by the predetermined offset; repeats the parsing ofthe length field until all the length fields contained in the headers ofas many first packets as the number of first packets identified by thecount field are completely parsed; and identifies a total length of thesecond packet by adding a header length of the second packet to thelengths of the IPv4 packets indicated by the parsed length fields.

Preferably, the packet type element identifies that the first packet isan IPv6 (Internet Protocol version 6) packet.

Preferably, the common protocol stack processor, if the C/S fieldindicates that a general IPv6 packet is contained in the payload of thesecond packet, parses a length field contained in a header of the firstpacket that is spaced apart from a payload start point of the secondpacket by a predetermined offset; and identifies a total length of thesecond packet by adding a sum of a header length of the first packet anda header length of the second packet to a length of the first packetindicated by the length field.

Preferably, if the C/S field indicates that a plurality of concatenatedIPv6 packets are contained in the payload of the second packet, theindicator part element further includes a count field indicating thenumber of first packets contained in the payload.

Preferably, the common protocol stack processor, if the C/S fieldindicates that a plurality of concatenated IPv6 packets are contained inthe payload of the second packet, parses a length field contained in aheader of the first packet that is spaced apart from a payload startpoint of the second packet by a predetermined offset, and parses alength field contained in a header of a next first packet that is spacedapart from a payload end point of the first packet identified by theparsed length field by the predetermined offset, repeats the parsing ofthe length field until all the length fields contained in the headers ofas many first packets as the number of first packets identified by thecount field are completely parsed; and identifies a total length of thesecond packet by adding not only a specific value obtained when a headerlength of the IPv6 packet is multiplied by the number of packetsidentified by the count field but also a header length of the secondpacket to the lengths of the IPv6 packets indicated by the parsed lengthfields.

Advantageous Effects

As is apparent from the above description, the broadcast receiveraccording to the embodiments can process packets of a link layer,irrespective of packet types received from an upper layer.

The broadcast receiver can use information contained in packets of ahigher layer of a protocol stack, prior to execution of higher layerprocessing, i.e., in the processing step of lower layer packets.

In accordance with the embodiments, the broadcast receiver can reduceoverhead encountered in a transmission process of higher layer packets.

DESCRIPTION OF DRAWINGS

The accompanying drawings, which are included to provide a furtherunderstanding of the invention and are incorporated in and constitute apart of this application, illustrate embodiment(s) of the invention andtogether with the description serve to explain the principle of theinvention. In the drawings:

FIG. 1 illustrates a structure of an apparatus for transmittingbroadcast signals for future broadcast services according to anembodiment of the present invention.

FIG. 2 illustrates an input formatting module according to oneembodiment of the present invention.

FIG. 3 illustrates an input formatting module according to anotherembodiment of the present invention.

FIG. 4 illustrates an input formatting module according to anotherembodiment of the present invention.

FIG. 5 illustrates a coding & modulation module according to anembodiment of the present invention.

FIG. 6 illustrates a frame structure module according to one embodimentof the present invention.

FIG. 7 illustrates a waveform generation module according to anembodiment of the present invention.

FIG. 8 illustrates a structure of an apparatus for receiving broadcastsignals for future broadcast services according to an embodiment of thepresent invention.

FIG. 9 illustrates a synchronization & demodulation module according toan embodiment of the present invention.

FIG. 10 illustrates a frame parsing module according to an embodiment ofthe present invention.

FIG. 11 illustrates a demapping & decoding module according to anembodiment of the present invention.

FIG. 12 illustrates an output processor according to an embodiment ofthe present invention.

FIG. 13 illustrates an output processor according to another embodimentof the present invention.

FIG. 14 illustrates a coding & modulation module according to anotherembodiment of the present invention.

FIG. 15 illustrates a demapping & decoding module according to anotherembodiment of the present invention.

FIG. 16 is a conceptual diagram illustrating combinations ofinterleavers on the condition that Signal Space Diversity (SSD) is notconsidered.

FIG. 17 shows the column-wise writing operations of the block timeinterleaver and the diagonal time interleaver according to the presentinvention.

FIG. 18 is a conceptual diagram illustrating a first scenario S2 fromamong combinations of the interleavers without consideration of a signalspace diversity (SSD).

FIG. 19 is a conceptual diagram of a second scenario S2 from amongcombinations of the interleavers without consideration of a signal spacediversity (SSD).

FIG. 20 is a conceptual diagram of a third scenario S3 from amongcombinations of the interleavers without consideration of signal spacediversity (SSD).

FIG. 21 is a conceptual diagram of a fourth scenario S4 from amongcombinations of the interleavers without consideration of a signal spacediversity (SSD).

FIG. 22 illustrates a structure of a random generator according to anembodiment of the present invention.

FIG. 23 illustrates a random generator according to an embodiment of thepresent invention.

FIG. 24 illustrates a random generator according to another embodimentof the present invention.

FIG. 25 illustrates a frequency interleaving process according to anembodiment of the present invention.

FIG. 26 is a conceptual diagram illustrating a frequency deinterleavingprocess according to an embodiment of the present invention.

FIG. 27 illustrates a frequency deinterleaving process according to anembodiment of the present invention.

FIG. 28 illustrates a process of generating a deinterleaved memory indexaccording to an embodiment of the present invention.

FIG. 29 illustrates a frequency interleaving process according to anembodiment of the present invention.

FIG. 30 illustrates a super-frame structure according to an embodimentof the present invention.

FIG. 31 illustrates a preamble insertion block according to anembodiment of the present invention.

FIG. 32 illustrates a preamble structure according to an embodiment ofthe present invention.

FIG. 33 illustrates a preamble detector according to an embodiment ofthe present invention.

FIG. 34 illustrates a correlation detector according to an embodiment ofthe present invention.

FIG. 35 shows graphs representing results obtained when the scramblingsequence according to an embodiment of the present invention is used.

FIG. 36 shows graphs representing results obtained when a scramblingsequence according to another embodiment of the present invention isused.

FIG. 37 shows graphs representing results obtained when a scramblingsequence according to another embodiment of the present invention isused.

FIG. 38 is a graph showing a result obtained when a scrambling sequenceaccording to another embodiment of the present invention is used.

FIG. 39 is a graph showing a result obtained when a scrambling sequenceaccording to another embodiment of the present invention is used.

FIG. 40 illustrates a signaling information interleaving procedureaccording to an embodiment of the present invention.

FIG. 41 illustrates a signaling information interleaving procedureaccording to another embodiment of the present invention.

FIG. 42 illustrates a signaling decoder according to an embodiment ofthe present invention.

FIG. 43 is a graph showing the performance of the signaling decoderaccording to an embodiment of the present invention.

FIG. 44 illustrates a preamble insertion block according to anotherembodiment of the present invention.

FIG. 45 illustrates a structure of signaling data in a preambleaccording to an embodiment of the present invention.

FIG. 46 illustrates a procedure of processing signaling data carried ona preamble according to one embodiment.

FIG. 47 illustrates a preamble structure repeated in the time domainaccording to one embodiment.

FIG. 48 illustrates a preamble detector and a correlation detectorincluded in the preamble detector according to an embodiment of thepresent invention.

FIG. 49 illustrates a preamble detector according to another embodimentof the present invention.

FIG. 50 illustrates a preamble detector and a signaling decoder includedin the preamble detector according to an embodiment of the presentinvention.

FIG. 51 is a view illustrating a frame structure of a broadcast systemaccording to an embodiment of the present invention.

FIG. 52 is a view illustrating DPs according to an embodiment of thepresent invention.

FIG. 53 is a view illustrating type1 DPs according to an embodiment ofthe present invention.

FIG. 54 is a view illustrating type2 DPs according to an embodiment ofthe present invention.

FIG. 55 is a view illustrating type3 DPs according to an embodiment ofthe present invention.

FIG. 56 is a view illustrating RBs according to an embodiment of thepresent invention.

FIG. 57 is a view illustrating a procedure for mapping RBs to framesaccording to an embodiment of the present invention.

FIG. 58 is a view illustrating RB mapping of type1 DPs according to anembodiment of the present invention.

FIG. 59 is a view illustrating RB mapping of type2 DPs according to anembodiment of the present invention.

FIG. 60 is a view illustrating RB mapping of type3 DPs according to anembodiment of the present invention.

FIG. 61 is a view illustrating RB mapping of type1 DPs according toanother embodiment of the present invention.

FIG. 62 is a view illustrating RB mapping of type1 DPs according toanother embodiment of the present invention.

FIG. 63 is a view illustrating RB mapping of type1 DPs according toanother embodiment of the present invention.

FIG. 64 is a view illustrating RB mapping of type2 DPs according toanother embodiment of the present invention.

FIG. 65 is a view illustrating RB mapping of type2 DPs according toanother embodiment of the present invention.

FIG. 66 is a view illustrating RB mapping of type3 DPs according toanother embodiment of the present invention.

FIG. 67 is a view illustrating RB mapping of type3 DPs according toanother embodiment of the present invention.

FIG. 68 is a view illustrating signaling information according to anembodiment of the present invention.

FIG. 69 is a graph showing the number of bits of a PLS according to thenumber of DPs according to an embodiment of the present invention.

FIG. 70 is a view illustrating a procedure for demapping DPs accordingto an embodiment of the present invention.

FIG. 71 is a view illustrating exemplary structures of three types ofmother codes applicable to perform LDPC encoding on PLS data in an FECencoder module according to another embodiment of the present invention.

FIG. 72 is a flowchart of a procedure for selecting a mother code typeused for LDPC encoding and determining the size of shortening accordingto another embodiment of the present invention.

FIG. 73 is a view illustrating a procedure for encoding adaptationparity according to another embodiment of the present invention.

FIG. 74 is a view illustrating a payload splitting mode for splittingPLS data input to the FEC encoder module before LDPC-encoding the inputPLS data according to another embodiment of the present invention.

FIG. 75 is a view illustrating a procedure for performing PLS repetitionand outputting a frame by the frame structure module 1200 according toanother embodiment of the present invention.

FIG. 76 is a view illustrating signal frame structures according toanother embodiment of the present invention.

FIG. 77 is a flowchart of a broadcast signal transmission methodaccording to another embodiment of the present invention.

FIG. 78 is a flowchart of a broadcast signal reception method accordingto another embodiment of the present invention.

FIG. 79 illustrates a waveform generation module and a synchronization &demodulation module according to another embodiment of the presentinvention.

FIG. 80 illustrates definition of a CP bearing SP and a CP not bearingSP according to an embodiment of the present invention.

FIG. 81 shows a reference index table according to an embodiment of thepresent invention.

FIG. 82 illustrates the concept of configuring a reference index tablein CP pattern generation method #1 using the position multiplexingmethod.

FIG. 83 illustrates a method for generating a reference index table inCP pattern generation method #1 using the position multiplexing methodaccording to an embodiment of the present invention.

FIG. 84 illustrates the concept of configuring a reference index tablein CP pattern generation method #2 using the position multiplexingmethod according to an embodiment of the present invention.

FIG. 85 illustrates a method for generating a reference index table inCP pattern generation method #2 using the position multiplexing method.

FIG. 86 illustrates a method for generating a reference index table inCP pattern generation method #3 using the position multiplexing methodaccording to an embodiment of the present invention.

FIG. 87 illustrates the concept of configuring a reference index tablein CP pattern generation method #1 using the pattern reversal method.

FIG. 88 illustrates a method for generating a reference index table inCP pattern generation method #1 using the pattern reversal methodaccording to an embodiment of the present invention.

FIG. 89 illustrates the concept of configuring a reference index tablein CP pattern generation method #2 using the pattern reversal methodaccording to an embodiment of the present invention.

FIG. 90 shows a table illustrating information related to a receptionmode according to an embodiment of the present invention.

FIG. 91 shows a bandwidth of the broadcast signal according to anembodiment of the present invention.

FIG. 92 shows tables including Tx parameters according to theembodiment.

FIG. 93 shows a table including Tx parameters capable of optimizing theeffective signal bandwidth (eBW) according to the embodiment.

FIG. 94 shows a table including Tx parameters for optimizing theeffective signal bandwidth (eBW) according to another embodiment of thepresent invention.

FIG. 95 shows a Table including Tx parameters for optimizing theeffective signal bandwidth (eBW) according to another embodiment of thepresent invention.

FIG. 96 shows Tx parameters according to another embodiment of thepresent invention.

FIG. 97 is a graph indicating Power Spectral Density (PSD) of atransmission (Tx) signal according to an embodiment of the presentinvention.

FIG. 98 is a table showing information related to the reception modeaccording to another embodiment of the present invention.

FIG. 99 shows the relationship between a maximum channel estimationrange and a guard interval according to the embodiment.

FIG. 100 shows a Table in which pilot parameters are defined accordingto an embodiment of the present invention.

FIG. 101 shows a Table in which pilot parameters of another embodimentare defined.

FIG. 102 shows the SISO pilot pattern according to an embodiment of thepresent invention.

FIG. 103 shows the MIXO-1 pilot pattern according to an embodiment ofthe present invention.

FIG. 104 shows the MIXO-2 pilot pattern according to an embodiment ofthe present invention.

FIG. 105 illustrates a MIMO encoding block diagram according to anembodiment of the present invention.

FIG. 106 shows a MIMO encoding scheme according to one embodiment of thepresent invention.

FIG. 107 is a diagram showing a PAM grid of an I or Q side according tonon-uniform QAM according to one embodiment of the present invention.

FIG. 108 is a diagram showing MIMO encoding input/output when the PH-eSMPI method is, applied to symbols mapped to non-uniform 64 QAM accordingto one embodiment of the present invention.

FIG. 109 is a graph for comparison in performance of MIMO encodingschemes according to the embodiment of the present invention.

FIG. 110 is a graph for comparison in performance of MIMO encodingschemes according to the embodiment of the present invention.

FIG. 111 is a graph for comparison in performance of MIMO encodingschemes according to the embodiment of the present invention.

FIG. 112 is a graph for comparison in performance of MIMO encodingschemes according to the embodiment of the present invention.

FIG. 113 is a diagram showing an embodiment of QAM-16 according to thepresent invention.

FIG. 114 is a diagram showing an embodiment of NUQ-64 for 5/15 code rateaccording to the present invention.

FIG. 115 is a diagram showing an embodiment of NUQ-64 for 6/15 code rateaccording to the present invention.

FIG. 116 is a diagram showing an embodiment of NUQ-64 for 7/15 code rateaccording to the present invention.

FIG. 117 is a diagram showing an embodiment of NUQ-64 for 8/15 code rateaccording to the present invention.

FIG. 118 is a diagram showing an embodiment of NUQ-64 for 9/15 and 10/15code rates according to the present invention.

FIG. 119 is a diagram showing an embodiment of NUQ-64 for 11/15 coderate according to the present invention.

FIG. 120 is a diagram showing an embodiment of NUQ-64 for 12/15 coderate according to the present invention.

FIG. 121 is a diagram showing an embodiment of NUQ-64 for 13/15 coderate according to the present invention.

FIG. 122 is a view illustrating a null packet deletion block 16000according to another embodiment of the present invention.

FIG. 123 is a view illustrating a null packet insertion block 17000according to another embodiment of the present invention.

FIG. 124 is a view illustrating a null packet spreading method accordingto an embodiment of the present invention.

FIG. 125 is a view illustrating a null packet offset method according toan embodiment of the present invention.

FIG. 126 is a flowchart illustrating a null packet spreading methodaccording to an embodiment of the present invention.

FIG. 127 is a conceptual diagram illustrating a protocol stack for thenext generation broadcast system based on hybrid according to anembodiment of the present invention.

FIG. 128 is a conceptual diagram illustrating an interface of a linklayer according to an embodiment of the present invention.

FIG. 129 is a conceptual diagram illustrating a packet structure of alink elayer according to an embodiment of the present invention.

FIG. 130 shows packet types dependent upon the packet type elementvalues according to an embodiment of the present invention.

FIG. 131 is a conceptual diagram illustrating a header structure of alink layer packet when an IP packet is transmitted to the link layeraccording to an embodiment of the present invention.

FIG. 132 is a conceptual diagram illustrating the meaning and headerstructures according to C/S field values.

FIG. 133 is a conceptual diagram illustrating the meaning according tothe count field values.

FIG. 134 is a conceptual diagram illustrating the meaning and segmentlengths according to values of Seg_Len_ID field.

FIG. 135 is a conceptual diagram illustrating an equation forencapsulating a normal packet and an equation for calculating a linklayer packet length.

FIG. 136 is a conceptual diagram illustrating a process forencapsulating a concatenated packet and an equation for calculating alink layer packet length.

FIG. 137 is a conceptual diagram illustrating a process for calculatingthe length of a concatenated packet including an IPv4 packet and anequation for calculating an offset value at which a length field of theIP packet is located.

FIG. 138 is a conceptual diagram illustrating a process for calculatingthe length of a concatenated packet including an IPv6 packet and anequation for calculating an offset value at which a length field of theIP packet is located.

FIG. 139 is a conceptual diagram illustrating an encapsulation processof a segmented packet according to an embodiment of the presentinvention.

FIG. 140 is a conceptual diagram illustrating a segmentation process ofan IP packet and header information of a link layer packet according toan embodiment of the present invention.

FIG. 141 is a conceptual diagram illustrating a segmentation process ofan IP packet including a cyclic redundancy check (CRC) according to anembodiment of the present invention.

FIG. 142 is a conceptual diagram illustrating a header structure of alink layer packet when MPEG-2 TS (Transport Stream) is input to a linklayer according to an embodiment of the present invention.

FIG. 143 shows the number of MPEG-2 TS packets contained in a payload ofthe link layer packet according to values of a count field.

FIG. 144 is a conceptual diagram illustrating a header of the MPEG-2 TSpacket according to an embodiment of the present invention.

FIG. 145 is a conceptual diagram illustrating a process for allowing atransceiver to change a usage of a transport error indicator fieldaccording to an embodiment of the present invention.

FIG. 146 is a conceptual diagram illustrating an encapsulation processof the MPEG-2 TS packet according to an embodiment of the presentinvention.

FIG. 147 is a conceptual diagram illustrating an encapsulation processof the MPEG-2 TS packet having the same PID according to an embodimentof the present invention.

FIG. 148 is a conceptual diagram illustrating an equation forcalculating the length of a link layer packet through a Common PIDreduction process and a Common PID reduction process.

FIG. 149 is a conceptual diagram illustrating the number of concatenatedMPEG-2 TS packets and the length of a link layer packet according tocount field values when Common PID reduction is used.

FIG. 150 is a conceptual diagram illustrating a process forencapsulating the MPEG-2 TS packet including a null packet according toan embodiment of the present invention.

FIG. 151 is a conceptual diagram illustrating a step for processing anindicator configured to count a removed null packet and an equation forcalculating the length of a link layer packet in the processing step.

FIG. 152 is a conceptual diagram illustrating a process forencapsulating the MPEG-2 TS packet including a null packet according toanother embodiment of the present invention.

FIG. 153 is a conceptual diagram illustrating a process forencapsulating the MPEG-2 TS packets including the same packetidentifiers (PIDs) in a stream including a null packet according to anembodiment of the present invention.

FIG. 154 is a conceptual diagram illustrating an equation forcalculating the length of a link layer packet when the MPEG-2 TS packetshaving the same PIDs are encapsulated in a stream including a nullpacket according to an embodiment of the present invention.

FIG. 155 is a conceptual diagram illustrating a link layer packetstructure for transmitting signaling information according to anembodiment of the present invention.

FIG. 156 is a conceptual diagram illustrating a link layer packetstructure for transmitting the framed packet according to an embodimentof the present invention.

FIG. 157 shows a syntax of the framed packet according to an embodimentof the present invention.

FIG. 158 is a block diagram illustrating a receiver of the nextgeneration broadcast system according to an embodiment of the presentinvention.

BEST MODE

Reference will now be made in detail to the preferred embodiments of thepresent invention, examples of which are illustrated in the accompanyingdrawings. The detailed description, which will be given below withreference to the accompanying drawings, is intended to explain exemplaryembodiments of the present invention, rather than to show the onlyembodiments that can be implemented according to the present invention.

Although most terms of elements in this specification have been selectedfrom general ones widely used in the art taking into considerationfunctions thereof in this specification, the terms may be changeddepending on the intention or convention of those skilled in the art orthe introduction of new technology. Some terms have been arbitrarilyselected by the applicant and their meanings are explained in thefollowing description as needed. Thus, the terms used in thisspecification should be construed based on the overall content of thisspecification together with the actual meanings of the terms rather thantheir simple names or meanings.

The term “signaling” in the present invention may indicate that serviceinformation (SI) that is transmitted and received from a broadcastsystem, an Internet system, and/or a broadcast/Internet convergencesystem. The service information (SI) may include broadcast serviceinformation (e.g., ATSC-SI and/or DVB-SI) received from the existingbroadcast systems.

The term “broadcast signal” may conceptually include not only signalsand/or data received from a terrestrial broadcast, a cable broadcast, asatellite broadcast, and/or a mobile broadcast, but also signals and/ordata received from bidirectional broadcast systems such as an Internetbroadcast, a broadband broadcast, a communication broadcast, a databroadcast, and/or VOD (Video On Demand).

The term “PLP” may indicate a predetermined unit for transmitting datacontained in a physical layer. Therefore, the term “PLP” may also bereplaced with the terms ‘data unit’ or ‘data pipe’ as necessary.

A hybrid broadcast service configured to interwork with the broadcastnetwork and/or the Internet network may be used as a representativeapplication to be used in a digital television (DTV) service. The hybridbroadcast service transmits, in real time, enhancement data related tobroadcast A/V (Audio/Video) contents transmitted through the terrestrialbroadcast network over the Internet, or transmits, in real time, someparts of the broadcast A/V contents over the Internet, such that userscan experience a variety of contents.

The present invention aims to provide a method for encapsulating an IPpacket, an MPEG-2 TS packet, and a packet applicable to other broadcastsystems in the next generation digital broadcast system in such a mannerthat the IP packet, the MPEG-2 TS packet, and the packet can betransmitted to a physical layer. In addition, the present inventionproposes a method for transmitting layer-2 signaling using the sameheader format.

The contents to be described hereinafter may be implemented by thedevice. For example, the following processes can be carried out by asignaling processor, a protocol processor, a processor, and/or a packetgenerator.

The present invention provides apparatuses and methods for transmittingand receiving broadcast signals for future broadcast services. Futurebroadcast services according to an embodiment of the present inventioninclude a terrestrial broadcast service, a mobile broadcast service, aUHDTV service, etc. The apparatuses and methods for transmittingaccording to an embodiment of the present invention may be categorizedinto a base profile for the terrestrial broadcast service, a handheldprofile for the mobile broadcast service and an advanced profile for theUHDTV service. In this case, the base profile can be used as a profilefor both the terrestrial broadcast service and the mobile broadcastservice. That is, the base profile can be used to define a concept of aprofile which includes the mobile profile. This can be changed accordingto intention of the designer.

The present invention may process broadcast signals for the futurebroadcast services through non-MIMO (Multiple Input Multiple Output) orMIMO according to one embodiment. A non-MIMO scheme according to anembodiment of the present invention may include a MISO (Multiple InputSingle Output) scheme, a SISO (Single Input Single Output) scheme, etc.

While MISO or MIMO uses two antennas in the following for convenience ofdescription, the present invention is applicable to systems using two ormore antennas.

FIG. 1 illustrates a structure of an apparatus for transmittingbroadcast signals for future broadcast services according to anembodiment of the present invention.

The apparatus for transmitting broadcast signals for future broadcastservices according to an embodiment of the present invention can includean input formatting module 1000, a coding & modulation module 1100, aframe structure module 1200, a waveform generation module 1300 and asignaling generation module 1400. A description will be given of theoperation of each module of the apparatus for transmitting broadcastsignals.

Referring to FIG. 1, the apparatus for transmitting broadcast signalsfor future broadcast services according to an embodiment of the presentinvention can receive MPEG-TSs, IP streams (v4/v6) and generic streams(GSs) as an input signal. In addition, the apparatus for transmittingbroadcast signals can receive management information about theconfiguration of each stream constituting the input signal and generatea final physical layer signal with reference to the received managementinformation.

The input formatting module 1000 according to an embodiment of thepresent invention can classify the input streams on the basis of astandard for coding and modulation or services or service components andoutput the input streams as a plurality of logical data pipes (or datapipes or DP data). The data pipe is a logical channel in the physicallayer that carries service data or related metadata, which may carry oneor multiple service(s) or service component(s). In addition, datatransmitted through each data pipe may be called DP data.

In addition, the input formatting module 1000 according to an embodimentof the present invention can divide each data pipe into blocks necessaryto perform coding and modulation and carry out processes necessary toincrease transmission efficiency or to perform scheduling. Details ofoperations of the input formatting module 1000 will be described later.

The coding & modulation module 1100 according to an embodiment of thepresent invention can perform forward error correction (FEC) encoding oneach data pipe received from the input formatting module 1000 such thatan apparatus for receiving broadcast signals can correct an error thatmay be generated on a transmission channel. In addition, the coding &modulation module 1100 according to an embodiment of the presentinvention can convert FEC output bit data to symbol data and interleavethe symbol data to correct burst error caused by a channel. As shown inFIG. 1, the coding & modulation module 1100 according to an embodimentof the present invention can divide the processed data such that thedivided data can be output through data paths for respective antennaoutputs in order to transmit the data through two or more Tx antennas.

The frame structure module 1200 according to an embodiment of thepresent invention can map the data output from the coding & modulationmodule 1100 to signal frames. The frame structure module 1200 accordingto an embodiment of the present invention can perform mapping usingscheduling information output from the input formatting module 1000 andinterleave data in the signal frames in order to obtain additionaldiversity gain.

The waveform generation module 1300 according to an embodiment of thepresent invention can convert the signal frames output from the framestructure module 1200 into a signal for transmission. In this case, thewaveform generation module 1300 according to an embodiment of thepresent invention can insert a preamble signal (or preamble) into thesignal for detection of the transmission apparatus and insert areference signal for estimating a transmission channel to compensate fordistortion into the signal. In addition, the waveform generation module1300 according to an embodiment of the present invention can provide aguard interval and insert a specific sequence into the same in order tooffset the influence of channel delay spread due to multi-pathreception. Additionally, the waveform generation module 1300 accordingto an embodiment of the present invention can perform a procedurenecessary for efficient transmission in consideration of signalcharacteristics such as a peak-to-average power ratio of the outputsignal.

The signaling generation module 1400 according to an embodiment of thepresent invention generates final physical layer signaling informationusing the input management information and information generated by theinput formatting module 1000, coding & modulation module 1100 and framestructure module 1200. Accordingly, a reception apparatus according toan embodiment of the present invention can decode a received signal bydecoding the signaling information.

As described above, the apparatus for transmitting broadcast signals forfuture broadcast services according to one embodiment of the presentinvention can provide terrestrial broadcast service, mobile broadcastservice, UHDTV service, etc. Accordingly, the apparatus for transmittingbroadcast signals for future broadcast services according to oneembodiment of the present invention can multiplex signals for differentservices in the time domain and transmit the same.

FIGS. 2, 3 and 4 illustrate the input formatting module 1000 accordingto embodiments of the present invention. A description will be given ofeach figure.

FIG. 2 illustrates an input formatting module according to oneembodiment of the present invention.

FIG. 2 shows an input formatting module when the input signal is asingle input stream.

Referring to FIG. 2, the input formatting module according to oneembodiment of the present invention can include a mode adaptation module2000 and a stream adaptation module 2100.

As shown in FIG. 2, the mode adaptation module 2000 can include an inputinterface block 2010, a CRC-8 encoder block 2020 and a BB headerinsertion block 2030. Description will be given of each block of themode adaptation module 2000.

The input interface block 2010 can divide the single input stream inputthereto into data pieces each having the length of a baseband (BB) frameused for FEC (BCH/LDPC) which will be performed later and output thedata pieces.

The CRC-8 encoder block 2020 can perform CRC encoding on BB frame datato add redundancy data thereto.

The BB header insertion block 2030 can insert, into the BB frame data, aheader including information such as mode adaptation type (TS/GS/IP), auser packet length, a data field length, user packet sync byte, startaddress of user packet sync byte in data field, a high efficiency modeindicator, an input stream synchronization field, etc.

As shown in FIG. 2, the stream adaptation module 2100 can include apadding insertion block 2110 and a BB scrambler block 2120. Descriptionwill be given of each block of the stream adaptation module 2100.

If data received from the mode adaptation module 2000 has a lengthshorter than an input data length necessary for FEC encoding, thepadding insertion block 2110 can insert a padding bit into the data suchthat the data has the input data length and output the data includingthe padding bit.

The BB scrambler block 2120 can randomize the input bit stream byperforming an XOR operation on the input bit stream and a pseudo randombinary sequence (PRBS).

The above-described blocks may be omitted or replaced by blocks havingsimilar or identical functions.

As shown in FIG. 2, the input formatting module can finally output datapipes to the coding & modulation module.

FIG. 3 illustrates an input formatting module according to anotherembodiment of the present invention.

FIG. 3 shows a mode adaptation module 3000 of the input formattingmodule when the input signal corresponds to multiple input streams.

The mode adaptation module 3000 of the input formatting module forprocessing the multiple input streams can independently process themultiple input streams.

Referring to FIG. 3, the mode adaptation module 3000 for respectivelyprocessing the multiple input streams can include input interfaceblocks, input stream synchronizer blocks 3100, compensating delay blocks3200, null packet deletion blocks 3300, CRC-8 encoder blocks and BBheader insertion blocks. Description will be given of each block of themode adaptation module 3000.

Operations of the input interface block, CRC-8 encoder block and BBheader insertion block correspond to those of the input interface block,CRC-8 encoder block and BB header insertion block described withreference to FIG. 2 and thus description thereof is omitted.

The input stream synchronizer block 3100 can transmit input stream clockreference (ISCR) information to generate timing information necessaryfor the apparatus for receiving broadcast signals to restore the TSs orGSs.

The compensating delay block 3200 can delay input data and output thedelayed input data such that the apparatus for receiving broadcastsignals can synchronize the input data if a delay is generated betweendata pipes according to processing of data including the timinginformation by the transmission apparatus.

The null packet deletion block 3300 can delete unnecessarily transmittedinput null packets from the input data, insert the number of deletednull packets into the input data based on positions in which the nullpackets are deleted and transmit the input data.

The above-described blocks may be omitted or replaced by blocks havingsimilar or identical functions.

FIG. 4 illustrates an input formatting module according to anotherembodiment of the present invention.

Specifically, FIG. 4 illustrates a stream adaptation module of the inputformatting module when the input signal corresponds to multiple inputstreams.

The stream adaptation module of the input formatting module when theinput signal corresponds to multiple input streams can include ascheduler 4000, a 1-frame delay block 4100, an in-band signaling orpadding insertion block 4200, a physical layer signaling generationblock 4300 and a BB scrambler block 4400. Description will be given ofeach block of the stream adaptation module.

The scheduler 4000 can perform scheduling for a MIMO system usingmultiple antennas having dual polarity. In addition, the scheduler 4000can generate parameters for use in signal processing blocks for antennapaths, such as a bit-to-cell demux block, a cell interleaver block, atime interleaver block, etc. included in the coding & modulation moduleillustrated in FIG. 1.

The 1-frame delay block 4100 can delay the input data by onetransmission frame such that scheduling information about the next framecan be transmitted through the current frame for in-band signalinginformation to be inserted into the data pipes.

The in-band signaling or padding insertion block 4200 can insertundelayed physical layer signaling (PLS)-dynamic signaling informationinto the data delayed by one transmission frame. In this case, thein-band signaling or padding insertion block 4200 can insert a paddingbit when a space for padding is present or insert in-band signalinginformation into the padding space. In addition, the scheduler 4000 canoutput physical layer signaling-dynamic signaling information about thecurrent frame separately from in-band signaling information.Accordingly, a cell mapper, which will be described later, can map inputcells according to scheduling information output from the scheduler4000.

The physical layer signaling generation block 4300 can generate physicallayer signaling data which will be transmitted through a preamble symbolof a transmission frame or spread and transmitted through a data symbolother than the in-band signaling information. In this case, the physicallayer signaling data according to an embodiment of the present inventioncan be referred to as signaling information. Furthermore, the physicallayer signaling data according to an embodiment of the present inventioncan be divided into PLS-pre information and PLS-post information. ThePLS-pre information can include parameters necessary to encode thePLS-post information and static PLS signaling data and the PLS-postinformation can include parameters necessary to encode the data pipes.The parameters necessary to encode the data pipes can be classified intostatic PLS signaling data and dynamic PLS signaling data. The static PLSsignaling data is a parameter commonly applicable to all frames includedin a super-frame and can be changed on a super-frame basis. The dynamicPLS signaling data is a parameter differently applicable to respectiveframes included in a super-frame and can be changed on a frame-by-framebasis. Accordingly, the reception apparatus can acquire the PLS-postinformation by decoding the PLS-pre information and decode desired datapipes by decoding the PLS-post information.

The BB scrambler block 4400 can generate a pseudo-random binary sequence(PRBS) and perform an XOR operation on the PRBS and the input bitstreams to decrease the peak-to-average power ratio (PAPR) of the outputsignal of the waveform generation block. As shown in FIG. 4, scramblingof the BB scrambler block 4400 is applicable to both data pipes andphysical layer signaling information.

The above-described blocks may be omitted or replaced by blocks havingsimilar or identical functions according to designer.

As shown in FIG. 4, the stream adaptation module can finally output thedata pipes to the coding & modulation module.

FIG. 5 illustrates a coding & modulation module according to anembodiment of the present invention.

The coding & modulation module shown in FIG. 5 corresponds to anembodiment of the coding & modulation module illustrated in FIG. 1.

As described above, the apparatus for transmitting broadcast signals forfuture broadcast services according to an embodiment of the presentinvention can provide a terrestrial broadcast service, mobile broadcastservice, UHDTV service, etc.

Since QoS (quality of service) depends on characteristics of a serviceprovided by the apparatus for transmitting broadcast signals for futurebroadcast services according to an embodiment of the present invention,data corresponding to respective services needs to be processed throughdifferent schemes. Accordingly, the coding & modulation module accordingto an embodiment of the present invention can independently process datapipes input thereto by independently applying SISO, MISO and MIMOschemes to the data pipes respectively corresponding to data paths.Consequently, the apparatus for transmitting broadcast signals forfuture broadcast services according to an embodiment of the presentinvention can control QoS for each service or service componenttransmitted through each data pipe.

Accordingly, the coding & modulation module according to an embodimentof the present invention can include a first block 5000 for SISO, asecond block 5100 for MISO, a third block 5200 for MIMO and a fourthblock 5300 for processing the PLS-pre/PLS-post information. The coding &modulation module illustrated in FIG. 5 is an exemplary and may includeonly the first block 5000 and the fourth block 5300, the second block5100 and the fourth block 5300 or the third block 5200 and the fourthblock 5300 according to design. That is, the coding & modulation modulecan include blocks for processing data pipes equally or differentlyaccording to design.

A description will be given of each block of the coding & modulationmodule.

The first block 5000 processes an input data pipe according to SISO andcan include an FEC encoder block 5010, a bit interleaver block 5020, abit-to-cell demux block 5030, a constellation mapper block 5040, a cellinterleaver block 5050 and a time interleaver block 5060.

The FEC encoder block 5010 can perform BCH encoding and LDPC encoding onthe input data pipe to add redundancy thereto such that the receptionapparatus can correct an error generated on a transmission channel.

The bit interleaver block 5020 can interleave bit streams of theFEC-encoded data pipe according to an interleaving rule such that thebit streams have robustness against burst error that may be generated onthe transmission channel. Accordingly, when deep fading or erasure isapplied to QAM symbols, errors can be prevented from being generated inconsecutive bits from among all codeword bits since interleaved bits aremapped to the QAM symbols.

The bit-to-cell demux block 5030 can determine the order of input bitstreams such that each bit in an FEC block can be transmitted withappropriate robustness in consideration of both the order of input bitstreams and a constellation mapping rule.

In addition, the bit interleaver block 5020 is located between the FECencoder block 5010 and the constellation mapper block 5040 and canconnect output bits of LDPC encoding performed by the FEC encoder block5010 to bit positions having different reliability values and optimalvalues of the constellation mapper in consideration of LDPC decoding ofthe apparatus for receiving broadcast signals. Accordingly, thebit-to-cell demux block 5030 can be replaced by a block having a similaror equal function.

The constellation mapper block 5040 can map a bit word input thereto toone constellation. In this case, the constellation mapper block 5040 canadditionally perform rotation & Q-delay. That is, the constellationmapper block 5040 can rotate input constellations according to arotation angle, divide the constellations into an in-phase component anda quadrature-phase component and delay only the quadrature-phasecomponent by an arbitrary value. Then, the constellation mapper block5040 can remap the constellations to new constellations using a pairedin-phase component and quadrature-phase component.

In addition, the constellation mapper block 5040 can move constellationpoints on a two-dimensional plane in order to find optimal constellationpoints. Through this process, capacity of the coding & modulation module1100 can be optimized. Furthermore, the constellation mapper block 5040can perform the above-described operation using IQ-balancedconstellation points and rotation. The constellation mapper block 5040can be replaced by a block having a similar or equal function.

The cell interleaver block 5050 can randomly interleave cellscorresponding to one FEC block and output the interleaved cells suchthat cells corresponding to respective FEC blocks can be output indifferent orders.

The time interleaver block 5060 can interleave cells belonging to aplurality of FEC blocks and output the interleaved cells. Accordingly,the cells corresponding to the FEC blocks are dispersed and transmittedin a period corresponding to a time interleaving depth and thusdiversity gain can be obtained.

The second block 5100 processes an input data pipe according to MISO andcan include the FEC encoder block, bit interleaver block, bit-to-celldemux block, constellation mapper block, cell interleaver block and timeinterleaver block in the same manner as the first block 5000. However,the second block 5100 is distinguished from the first block 5000 in thatthe second block 5100 further includes a MISO processing block 5110. Thesecond block 5100 performs the same procedure including the inputoperation to the time interleaver operation as those of the first block5000 and thus description of the corresponding blocks is omitted.

The MISO processing block 5110 can encode input cells according to aMISO encoding matrix providing transmit diversity and outputMISO-processed data through two paths. MISO processing according to oneembodiment of the present invention can include OSTBC (orthogonal spacetime block coding)/OSFBC (orthogonal space frequency block coding,Alamouti coding).

The third block 5200 processes an input data pipe according to MIMO andcan include the FEC encoder block, bit interleaver block, bit-to-celldemux block, constellation mapper block, cell interleaver block and timeinterleaver block in the same manner as the second block 5100, as shownin FIG. 5. However, the data processing procedure of the third block5200 is different from that of the second block 5100 since the thirdblock 5200 includes a MIMO processing block 5220.

That is, in the third block 5200, basic roles of the FEC encoder blockand the bit interleaver block are identical to those of the first andsecond blocks 5000 and 5100 although functions thereof may be differentfrom those of the first and second blocks 5000 and 5100.

The bit-to-cell demux block 5210 can generate as many output bit streamsas input bit streams of MIMO processing and output the output bitstreams through MIMO paths for MIMO processing. In this case, thebit-to-cell demux block 5210 can be designed to optimize the decodingperformance of the reception apparatus in consideration ofcharacteristics of LDPC and MIMO processing.

Basic roles of the constellation mapper block, cell interleaver blockand time interleaver block are identical to those of the first andsecond blocks 5000 and 5100 although functions thereof may be differentfrom those of the first and second blocks 5000 and 5100. As shown inFIG. 5, as many constellation mapper blocks, cell interleaver blocks andtime interleaver blocks as the number of MIMO paths for MIMO processingcan be present. In this case, the constellation mapper blocks, cellinterleaver blocks and time interleaver blocks can operate equally orindependently for data input through the respective paths.

The MIMO processing block 5220 can perform MIMO processing on two inputcells using a MIMO encoding matrix and output the MIMO-processed datathrough two paths. The MIMO encoding matrix according to an embodimentof the present invention can include spatial multiplexing, Golden code,full-rate full diversity code, linear dispersion code, etc.

The fourth block 5300 processes the PLS-pre/PLS-post information and canperform SISO or MISO processing.

The basic roles of the bit interleaver block, bit-to-cell demux block,constellation mapper block, cell interleaver block, time interleaverblock and MISO processing block included in the fourth block 5300correspond to those of the second block 5100 although functions thereofmay be different from those of the second block 5100.

A shortened/punctured FEC encoder block 5310 included in the fourthblock 5300 can process PLS data using an FEC encoding scheme for a PLSpath provided for a case in which the length of input data is shorterthan a length necessary to perform FEC encoding.

Specifically, the shortened/punctured FEC encoder block 5310 can performBCH encoding on input bit streams, pad 0s corresponding to a desiredinput bit stream length necessary for normal LDPC encoding, carry outLDPC encoding and then remove the padded 0s to puncture parity bits suchthat an effective code rate becomes equal to or lower than the data piperate.

The blocks included in the first block 5000 to fourth block 5300 may beomitted or replaced by blocks having similar or identical functionsaccording to design.

As illustrated in FIG. 5, the coding & modulation module can output thedata pipes (or DP data), PLS-pre information and PLS-post informationprocessed for the respective paths to the frame structure module.

FIG. 6 illustrates a frame structure module according to one embodimentof the present invention.

The frame structure module shown in FIG. 6 corresponds to an embodimentof the frame structure module 1200 illustrated in FIG. 1.

The frame structure module according to one embodiment of the presentinvention can include at least one cell-mapper 6000, at least one delaycompensation module 6100 and at least one block interleaver 6200. Thenumber of cell mappers 6000, delay compensation modules 6100 and blockinterleavers 6200 can be changed. A description will be given of eachmodule of the frame structure block.

The cell-mapper 6000 can allocate cells corresponding to SISO-, MISO- orMIMO-processed data pipes output from the coding & modulation module,cells corresponding to common data commonly applicable to the data pipesand cells corresponding to the PLS-pre/PLS-post information to signalframes according to scheduling information. The common data refers tosignaling information commonly applied to all or some data pipes and canbe transmitted through a specific data pipe. The data pipe through whichthe common data is transmitted can be referred to as a common data pipeand can be changed according to design.

When the apparatus for transmitting broadcast signals according to anembodiment of the present invention uses two output antennas andAlamouti coding is used for MISO processing, the cell-mapper 6000 canperform pair-wise cell mapping in order to maintain orthogonalityaccording to Alamouti encoding. That is, the cell-mapper 6000 canprocess two consecutive cells of the input cells as one unit and map theunit to a frame. Accordingly, paired cells in an input pathcorresponding to an output path of each antenna can be allocated toneighboring positions in a transmission frame.

The delay compensation block 6100 can obtain PLS data corresponding tothe current transmission frame by delaying input PLS data cells for thenext transmission frame by one frame. In this case, the PLS datacorresponding to the current frame can be transmitted through a preamblepart in the current signal frame and PLS data corresponding to the nextsignal frame can be transmitted through a preamble part in the currentsignal frame or in-band signaling in each data pipe of the currentsignal frame. This can be changed by the designer.

The block interleaver 6200 can obtain additional diversity gain byinterleaving cells in a transport block corresponding to the unit of asignal frame. In addition, the block interleaver 6200 can performinterleaving by processing two consecutive cells of the input cells asone unit when the above-described pair-wise cell mapping is performed.Accordingly, cells output from the block interleaver 6200 can be twoconsecutive identical cells.

When pair-wise mapping and pair-wise interleaving are performed, atleast one cell mapper and at least one block interleaver can operateequally or independently for data input through the paths.

The above-described blocks may be omitted or replaced by blocks havingsimilar or identical functions according to design.

As illustrated in FIG. 6, the frame structure module can output at leastone signal frame to the waveform generation module.

FIG. 7 illustrates a waveform generation module according to anembodiment of the present invention.

The waveform generation module illustrated in FIG. 7 corresponds to anembodiment of the waveform generation module 1300 described withreference to FIG. 1.

The waveform generation module according to an embodiment of the presentinvention can modulate and transmit as many signal frames as the numberof antennas for receiving and outputting signal frames output from theframe structure module illustrated in FIG. 6.

Specifically, the waveform generation module illustrated in FIG. 7 is anembodiment of a waveform generation module of an apparatus fortransmitting broadcast signals using m Tx antennas and can include mprocessing blocks for modulating and outputting frames corresponding tom paths. The m processing blocks can perform the same processingprocedure. A description will be given of operation of the firstprocessing block 7000 from among the m processing blocks.

The first processing block 7000 can include a reference signal & PAPRreduction block 7100, an inverse waveform transform block 7200, a PAPRreduction in time block 7300, a guard sequence insertion block 7400, apreamble insertion block 7500, a waveform processing block 7600, othersystem insertion block 7700 and a DAC (digital analog converter) block7800.

The reference signal insertion & PAPR reduction block 7100 can insert areference signal into a predetermined position of each signal block andapply a PAPR reduction scheme to reduce a PAPR in the time domain. If abroadcast transmission/reception system according to an embodiment ofthe present invention corresponds to an OFDM system, the referencesignal insertion & PAPR reduction block 7100 can use a method ofreserving some active subcarriers rather than using the same. Inaddition, the reference signal insertion & PAPR reduction block 7100 maynot use the PAPR reduction scheme as an optional feature according tobroadcast transmission/reception system.

The inverse waveform transform block 7200 can transform an input signalin a manner of improving transmission efficiency and flexibility inconsideration of transmission channel characteristics and systemarchitecture. If the broadcast transmission/reception system accordingto an embodiment of the present invention corresponds to an OFDM system,the inverse waveform transform block 7200 can employ a method oftransforming a frequency domain signal into a time domain signal throughinverse FFT operation. If the broadcast transmission/reception systemaccording to an embodiment of the present invention corresponds to asingle carrier system, the inverse waveform transform block 7200 may notbe used in the waveform generation module.

The PAPR reduction in time block 7300 can use a method for reducing PAPRof an input signal in the time domain. If the broadcasttransmission/reception system according to an embodiment of the presentinvention corresponds to an OFDM system, the PAPR reduction in timeblock 7300 may use a method of simply clipping peak amplitude.Furthermore, the PAPR reduction in time block 7300 may not be used inthe broadcast transmission/reception system according to an embodimentof the present invention since it is an optional feature.

The guard sequence insertion block 7400 can provide a guard intervalbetween neighboring signal blocks and insert a specific sequence intothe guard interval as necessary in order to minimize the influence ofdelay spread of a transmission channel. Accordingly, the receptionapparatus can easily perform synchronization or channel estimation. Ifthe broadcast transmission/reception system according to an embodimentof the present invention corresponds to an OFDM system, the guardsequence insertion block 7400 may insert a cyclic prefix into a guardinterval of an OFDM symbol.

The preamble insertion block 7500 can insert a signal of a known type(e.g. the preamble or preamble symbol) agreed upon between thetransmission apparatus and the reception apparatus into a transmissionsignal such that the reception apparatus can rapidly and efficientlydetect a target system signal. If the broadcast transmission/receptionsystem according to an embodiment of the present invention correspondsto an OFDM system, the preamble insertion block 7500 can define a signalframe composed of a plurality of OFDM symbols and insert a preamblesymbol into the beginning of each signal frame. That is, the preamblecarries basic PLS data and is located in the beginning of a signalframe.

The waveform processing block 7600 can perform waveform processing on aninput baseband signal such that the input baseband signal meets channeltransmission characteristics. The waveform processing block 7600 may usea method of performing square-root-raised cosine (SRRC) filtering toobtain a standard for out-of-band emission of a transmission signal. Ifthe broadcast transmission/reception system according to an embodimentof the present invention corresponds to a multi-carrier system, thewaveform processing block 7600 may not be used.

The other system insertion block 7700 can multiplex signals of aplurality of broadcast transmission/reception systems in the time domainsuch that data of two or more different broadcast transmission/receptionsystems providing broadcast services can be simultaneously transmittedin the same RF signal bandwidth. In this case, the two or more differentbroadcast transmission/reception systems refer to systems providingdifferent broadcast services. The different broadcast services may referto a terrestrial broadcast service, mobile broadcast service, etc. Datarelated to respective broadcast services can be transmitted throughdifferent frames.

The DAC block 7800 can convert an input digital signal into an analogsignal and output the analog signal. The signal output from the DACblock 7800 can be transmitted through m output antennas. A Tx antennaaccording to an embodiment of the present invention can have vertical orhorizontal polarity.

The above-described blocks may be omitted or replaced by blocks havingsimilar or identical functions according to design.

FIG. 8 illustrates a structure of an apparatus for receiving broadcastsignals for future broadcast services according to an embodiment of thepresent invention.

The apparatus for receiving broadcast signals for future broadcastservices according to an embodiment of the present invention cancorrespond to the apparatus for transmitting broadcast signals forfuture broadcast services, described with reference to FIG. 1. Theapparatus for receiving broadcast signals for future broadcast servicesaccording to an embodiment of the present invention can include asynchronization & demodulation module 8000, a frame parsing module 8100,a demapping & decoding module 8200, an output processor 8300 and asignaling decoding module 8400. A description will be given of operationof each module of the apparatus for receiving broadcast signals.

The synchronization & demodulation module 8000 can receive input signalsthrough m Rx antennas, perform signal detection and synchronization withrespect to a system corresponding to the apparatus for receivingbroadcast signals and carry out demodulation corresponding to a reverseprocedure of the procedure performed by the apparatus for transmittingbroadcast signals.

The frame parsing module 8100 can parse input signal frames and extractdata through which a service selected by a user is transmitted. If theapparatus for transmitting broadcast signals performs interleaving, theframe parsing module 8100 can carry out deinterleaving corresponding toa reverse procedure of interleaving. In this case, the positions of asignal and data that need to be extracted can be obtained by decodingdata output from the signaling decoding module 8400 to restorescheduling information generated by the apparatus for transmittingbroadcast signals.

The demapping & decoding module 8200 can convert the input signals intobit domain data and then deinterleave the same as necessary. Thedemapping & decoding module 8200 can perform demapping for mappingapplied for transmission efficiency and correct an error generated on atransmission channel through decoding. In this case, the demapping &decoding module 8200 can obtain transmission parameters necessary fordemapping and decoding by decoding the data output from the signalingdecoding module 8400.

The output processor 8300 can perform reverse procedures of variouscompression/signal processing procedures which are applied by theapparatus for transmitting broadcast signals to improve transmissionefficiency. In this case, the output processor 8300 can acquirenecessary control information from data output from the signalingdecoding module 8400. The output of the output processor 8300corresponds to a signal input to the apparatus for transmittingbroadcast signals and may be MPEG-TSs, IP streams (v4 or v6) and genericstreams.

The signaling decoding module 8400 can obtain PLS information from thesignal demodulated by the synchronization & demodulation module 8000. Asdescribed above, the frame parsing module 8100, demapping & decodingmodule 8200 and output processor 8300 can execute functions thereofusing the data output from the signaling decoding module 8400.

FIG. 9 illustrates a synchronization & demodulation module according toan embodiment of the present invention.

The synchronization & demodulation module shown in FIG. 9 corresponds toan embodiment of the synchronization & demodulation module describedwith reference to FIG. 8. The synchronization & demodulation moduleshown in FIG. 9 can perform a reverse operation of the operation of thewaveform generation module illustrated in FIG. 7.

As shown in FIG. 9, the synchronization & demodulation module accordingto an embodiment of the present invention corresponds to asynchronization & demodulation module of an apparatus for receivingbroadcast signals using m Rx antennas and can include m processingblocks for demodulating signals respectively input through m paths. Them processing blocks can perform the same processing procedure. Adescription will be given of operation of the first processing block9000 from among the m processing blocks.

The first processing block 9000 can include a tuner 9100, an ADC block9200, a preamble detector 9300, a guard sequence detector 9400, awaveform transform block 9500, a time/frequency synchronization block9600, a reference signal detector 9700, a channel equalizer 9800 and aninverse waveform transform block 9900.

The tuner 9100 can select a desired frequency band, compensate for themagnitude of a received signal and output the compensated signal to theADC block 9200.

The ADC block 9200 can convert the signal output from the tuner 9100into a digital signal.

The preamble detector 9300 can detect a preamble (or preamble signal orpreamble symbol) in order to check whether or not the digital signal isa signal of the system corresponding to the apparatus for receivingbroadcast signals. In this case, the preamble detector 9300 can decodebasic transmission parameters received through the preamble.

The guard sequence detector 9400 can detect a guard sequence in thedigital signal. The time/frequency synchronization block 9600 canperform time/frequency synchronization using the detected guard sequenceand the channel equalizer 9800 can estimate a channel through areceived/restored sequence using the detected guard sequence.

The waveform transform block 9500 can perform a reverse operation ofinverse waveform transform when the apparatus for transmitting broadcastsignals has performed inverse waveform transform. When the broadcasttransmission/reception system according to one embodiment of the presentinvention is a multi-carrier system, the waveform transform block 9500can perform FFT. Furthermore, when the broadcast transmission/receptionsystem according to an embodiment of the present invention is a singlecarrier system, the waveform transform block 9500 may not be used if areceived time domain signal is processed in the frequency domain orprocessed in the time domain.

The time/frequency synchronization block 9600 can receive output data ofthe preamble detector 9300, guard sequence detector 9400 and referencesignal detector 9700 and perform time synchronization and carrierfrequency synchronization including guard sequence detection and blockwindow positioning on a detected signal. Here, the time/frequencysynchronization block 9600 can feed back the output signal of thewaveform transform block 9500 for frequency synchronization.

The reference signal detector 9700 can detect a received referencesignal.

Accordingly, the apparatus for receiving broadcast signals according toan embodiment of the present invention can perform synchronization orchannel estimation.

The channel equalizer 9800 can estimate a transmission channel from eachTx antenna to each Rx antenna from the guard sequence or referencesignal and perform channel equalization for received data using theestimated channel.

The inverse waveform transform block 9900 may restore the originalreceived data domain when the waveform transform block 9500 performswaveform transform for efficient synchronization and channelestimation/equalization. If the broadcast transmission/reception systemaccording to an embodiment of the present invention is a single carriersystem, the waveform transform block 9500 can perform FFT in order tocarry out synchronization/channel estimation/equalization in thefrequency domain and the inverse waveform transform block 9900 canperform IFFT on the channel-equalized signal to restore transmitted datasymbols. If the broadcast transmission/reception system according to anembodiment of the present invention is a multi-carrier system, theinverse waveform transform block 9900 may not be used.

The above-described blocks may be omitted or replaced by blocks havingsimilar or identical functions according to design.

FIG. 10 illustrates a frame parsing module according to an embodiment ofthe present invention.

The frame parsing module illustrated in FIG. 10 corresponds to anembodiment of the frame parsing module described with reference to FIG.8. The frame parsing module shown in FIG. 10 can perform a reverseoperation of the operation of the frame structure module illustrated inFIG. 6.

As shown in FIG. 10, the frame parsing module according to an embodimentof the present invention can include at least one block deinterleaver10000 and at least one cell demapper 10100.

The block deinterleaver 10000 can deinterleave data input through datapaths of the m Rx antennas and processed by the synchronization &demodulation module on a signal block basis. In this case, if theapparatus for transmitting broadcast signals performs pair-wiseinterleaving as illustrated in FIG. 8, the block deinterleaver 10000 canprocess two consecutive pieces of data as a pair for each input path.Accordingly, the block interleaver 10000 can output two consecutivepieces of data even when deinterleaving has been performed. Furthermore,the block deinterleaver 10000 can perform a reverse operation of theinterleaving operation performed by the apparatus for transmittingbroadcast signals to output data in the original order.

The cell demapper 10100 can extract cells corresponding to common data,cells corresponding to data pipes and cells corresponding to PLS datafrom received signal frames. The cell demapper 10100 can merge datadistributed and transmitted and output the same as a stream asnecessary. When two consecutive pieces of cell input data are processedas a pair and mapped in the apparatus for transmitting broadcastsignals, as shown in FIG. 6, the cell demapper 10100 can performpair-wise cell demapping for processing two consecutive input cells asone unit as a reverse procedure of the mapping operation of theapparatus for transmitting broadcast signals.

In addition, the cell demapper 10100 can extract PLS signaling datareceived through the current frame as PLS-pre & PLS-post data and outputthe PLS-pre & PLS-post data.

The above-described blocks may be omitted or replaced by blocks havingsimilar or identical functions according to design.

FIG. 11 illustrates a demapping & decoding module according to anembodiment of the present invention.

The demapping & decoding module shown in FIG. 11 corresponds to anembodiment of the demapping & decoding module illustrated in FIG. 8. Thedemapping & decoding module shown in FIG. 11 can perform a reverseoperation of the operation of the coding & modulation module illustratedin FIG. 5.

The coding & modulation module of the apparatus for transmittingbroadcast signals according to an embodiment of the present inventioncan process input data pipes by independently applying SISO, MISO andMIMO thereto for respective paths, as described above. Accordingly, thedemapping & decoding module illustrated in FIG. 11 can include blocksfor processing data output from the frame parsing module according toSISO, MISO and MIMO in response to the apparatus for transmittingbroadcast signals.

As shown in FIG. 11, the demapping & decoding module according to anembodiment of the present invention can include a first block 11000 forSISO, a second block 11100 for MISO, a third block 11200 for MIMO and afourth block 11300 for processing the PLS-pre/PLS-post information. Thedemapping & decoding module shown in FIG. 11 is exemplary and mayinclude only the first block 11000 and the fourth block 11300, only thesecond block 11100 and the fourth block 11300 or only the third block11200 and the fourth block 11300 according to design. That is, thedemapping & decoding module can include blocks for processing data pipesequally or differently according to design.

A description will be given of each block of the demapping & decodingmodule.

The first block 11000 processes an input data pipe according to SISO andcan include a time deinterleaver block 11010, a cell deinterleaver block11020, a constellation demapper block 11030, a cell-to-bit mux block11040, a bit deinterleaver block 11050 and an FEC decoder block 11060.

The time deinterleaver block 11010 can perform a reverse process of theprocess performed by the time interleaver block 5060 illustrated in FIG.5. That is, the time deinterleaver block 11010 can deinterleave inputsymbols interleaved in the time domain into original positions thereof.

The cell deinterleaver block 11020 can perform a reverse process of theprocess performed by the cell interleaver block 5050 illustrated in FIG.5. That is, the cell deinterleaver block 11020 can deinterleavepositions of cells spread in one FEC block into original positionsthereof.

The constellation demapper block 11030 can perform a reverse process ofthe process performed by the constellation mapper block 5040 illustratedin FIG. 5. That is, the constellation demapper block 11030 can demap asymbol domain input signal to bit domain data. In addition, theconstellation demapper block 11030 may perform hard decision and outputdecided bit data. Furthermore, the constellation demapper block 11030may output a log-likelihood ratio (LLR) of each bit, which correspondsto a soft decision value or probability value. If the apparatus fortransmitting broadcast signals applies a rotated constellation in orderto obtain additional diversity gain, the constellation demapper block11030 can perform 2-dimensional LLR demapping corresponding to therotated constellation. Here, the constellation demapper block 11030 cancalculate the LLR such that a delay applied by the apparatus fortransmitting broadcast signals to the I or Q component can becompensated.

The cell-to-bit mux block 11040 can perform a reverse process of theprocess performed by the bit-to-cell demux block 5030 illustrated inFIG. 5. That is, the cell-to-bit mux block 11040 can restore bit datamapped by the bit-to-cell demux block 5030 to the original bit streams.

The bit deinterleaver block 11050 can perform a reverse process of theprocess performed by the bit interleaver 5020 illustrated in FIG. 5.That is, the bit deinterleaver block 11050 can deinterleave the bitstreams output from the cell-to-bit mux block 11040 in the originalorder.

The FEC decoder block 11060 can perform a reverse process of the processperformed by the FEC encoder block 5010 illustrated in FIG. 5. That is,the FEC decoder block 11060 can correct an error generated on atransmission channel by performing LDPC decoding and BCH decoding.

The second block 11100 processes an input data pipe according to MISOand can include the time deinterleaver block, cell deinterleaver block,constellation demapper block, cell-to-bit mux block, bit deinterleaverblock and FEC decoder block in the same manner as the first block 11000,as shown in FIG. 11. However, the second block 11100 is distinguishedfrom the first block 11000 in that the second block 11100 furtherincludes a MISO decoding block 11110. The second block 11100 performsthe same procedure including time deinterleaving operation to outputtingoperation as the first block 11000 and thus description of thecorresponding blocks is omitted.

The MISO decoding block 11110 can perform a reverse operation of theoperation of the MISO processing block 5110 illustrated in FIG. 5. Ifthe broadcast transmission/reception system according to an embodimentof the present invention uses STBC, the MISO decoding block 11110 canperform Alamouti decoding.

The third block 11200 processes an input data pipe according to MIMO andcan include the time deinterleaver block, cell deinterleaver block,constellation demapper block, cell-to-bit mux block, bit deinterleaverblock and FEC decoder block in the same manner as the second block11100, as shown in FIG. 11. However, the third block 11200 isdistinguished from the second block 11100 in that the third block 11200further includes a MIMO decoding block 11210. The basic roles of thetime deinterleaver block, cell deinterleaver block, constellationdemapper block, cell-to-bit mux block and bit deinterleaver blockincluded in the third block 11200 are identical to those of thecorresponding blocks included in the first and second blocks 11000 and11100 although functions thereof may be different from the first andsecond blocks 11000 and 11100.

The MIMO decoding block 11210 can receive output data of the celldeinterleaver for input signals of the m Rx antennas and perform MIMOdecoding as a reverse operation of the operation of the MIMO processingblock 5220 illustrated in FIG. 5. The MIMO decoding block 11210 canperform maximum likelihood decoding to obtain optimal decodingperformance or carry out sphere decoding with reduced complexity.Otherwise, the MIMO decoding block 11210 can achieve improved decodingperformance by performing MMSE detection or carrying out iterativedecoding with MMSE detection.

The fourth block 11300 processes the PLS-pre/PLS-post information andcan perform SISO or MISO decoding. The fourth block 11300 can carry outa reverse process of the process performed by the fourth block 5300described with reference to FIG. 5.

The basic roles of the time deinterleaver block, cell deinterleaverblock, constellation demapper block, cell-to-bit mux block and bitdeinterleaver block included in the fourth block 11300 are identical tothose of the corresponding blocks of the first, second and third blocks11000, 11100 and 11200 although functions thereof may be different fromthe first, second and third blocks 11000, 11100 and 11200.

The shortened/punctured FEC decoder 11310 included in the fourth block11300 can perform a reverse process of the process performed by theshortened/punctured FEC encoder block 5310 described with reference toFIG. 5. That is, the shortened/punctured FEC decoder 11310 can performde-shortening and de-puncturing on data shortened/punctured according toPLS data length and then carry out FEC decoding thereon. In this case,the FEC decoder used for data pipes can also be used for PLS.Accordingly, additional FEC decoder hardware for the PLS only is notneeded and thus system design is simplified and efficient coding isachieved.

The above-described blocks may be omitted or replaced by blocks havingsimilar or identical functions according to design.

The demapping & decoding module according to an embodiment of thepresent invention can output data pipes and PLS information processedfor the respective paths to the output processor, as illustrated in FIG.11.

FIGS. 12 and 13 illustrate output processors according to embodiments ofthe present invention.

FIG. 12 illustrates an output processor according to an embodiment ofthe present invention.

The output processor illustrated in FIG. 12 corresponds to an embodimentof the output processor illustrated in FIG. 8. The output processorillustrated in FIG. 12 receives a single data pipe output from thedemapping & decoding module and outputs a single output stream. Theoutput processor can perform a reverse operation of the operation of theinput formatting module illustrated in FIG. 2.

The output processor shown in FIG. 12 can include a BB scrambler block12000, a padding removal block 12100, a CRC-8 decoder block 12200 and aBB frame processor block 12300.

The BB scrambler block 12000 can descramble an input bit stream bygenerating the same PRBS as that used in the apparatus for transmittingbroadcast signals for the input bit stream and carrying out an XORoperation on the PRBS and the bit stream.

The padding removal block 12100 can remove padding bits inserted by theapparatus for transmitting broadcast signals as necessary.

The CRC-8 decoder block 12200 can check a block error by performing CRCdecoding on the bit stream received from the padding removal block12100.

The BB frame processor block 12300 can decode information transmittedthrough a BB frame header and restore MPEG-TSs, IP streams (v4 or v6) orgeneric streams using the decoded information.

The above-described blocks may be omitted or replaced by blocks havingsimilar or identical functions according to design.

FIG. 13 illustrates an output processor according to another embodimentof the present invention.

The output processor shown in FIG. 13 corresponds to an embodiment ofthe output processor illustrated in FIG. 8. The output processor shownin FIG. 13 receives multiple data pipes output from the demapping &decoding module. Decoding multiple data pipes can include a process ofmerging common data commonly applicable to a plurality of data pipes anddata pipes related thereto and decoding the same or a process ofsimultaneously decoding a plurality of services or service components(including a scalable video service) by the apparatus for receivingbroadcast signals.

The output processor shown in FIG. 13 can include a BB descramblerblock, a padding removal block, a CRC-8 decoder block and a BB frameprocessor block as the output processor illustrated in FIG. 12. Thebasic roles of these blocks correspond to those of the blocks describedwith reference to FIG. 12 although operations thereof may differ fromthose of the blocks illustrated in FIG. 12.

A de-jitter buffer block 13000 included in the output processor shown inFIG. 13 can compensate for a delay, inserted by the apparatus fortransmitting broadcast signals for synchronization of multiple datapipes, according to a restored TTO (time to output) parameter.

A null packet insertion block 13100 can restore a null packet removedfrom a stream with reference to a restored DNP (deleted null packet) andoutput common data.

A TS clock regeneration block 13200 can restore time synchronization ofoutput packets based on ISCR (input stream time reference) information.

A TS recombining block 13300 can recombine the common data and datapipes related thereto, output from the null packet insertion block13100, to restore the original MPEG-TSs, IP streams (v4 or v6) orgeneric streams. The TTO, DNT and ISCR information can be obtainedthrough the BB frame header.

An in-band signaling decoding block 13400 can decode and output in-bandphysical layer signaling information transmitted through a padding bitfield in each FEC frame of a data pipe.

The output processor shown in FIG. 13 can BB-descramble the PLS-preinformation and PLS-post information respectively input through aPLS-pre path and a PLS-post path and decode the descrambled data torestore the original PLS data. The restored PLS data is delivered to asystem controller included in the apparatus for receiving broadcastsignals. The system controller can provide parameters necessary for thesynchronization & demodulation module, frame parsing module, demapping &decoding module and output processor module of the apparatus forreceiving broadcast signals.

The above-described blocks may be omitted or replaced by blocks havingsimilar r identical functions according to design.

FIG. 14 illustrates a coding & modulation module according to anotherembodiment of the present invention.

The coding & modulation module shown in FIG. 14 corresponds to anotherembodiment of the coding & modulation module illustrated in FIGS. 1 to5.

To control QoS for each service or service component transmitted througheach data pipe, as described above with reference to FIG. 5, the coding& modulation module shown in FIG. 14 can include a first block 14000 forSISO, a second block 14100 for MISO, a third block 14200 for MIMO and afourth block 14300 for processing the PLS-pre/PLS-post information. Inaddition, the coding & modulation module can include blocks forprocessing data pipes equally or differently according to the design.The first to fourth blocks 14000 to 14300 shown in FIG. 14 are similarto the first to fourth blocks 5000 to 5300 illustrated in FIG. 5.

However, the first to fourth blocks 14000 to 14300 shown in FIG. 14 aredistinguished from the first to fourth blocks 5000 to 5300 illustratedin FIG. 5 in that a constellation mapper 14010 included in the first tofourth blocks 14000 to 14300 has a function different from the first tofourth blocks 5000 to 5300 illustrated in FIG. 5, a rotation & I/Qinterleaver block 14020 is present between the cell interleaver and thetime interleaver of the first to fourth blocks 14000 to 14300illustrated in FIG. 14 and the third block 14200 for MIMO has aconfiguration different from the third block 5200 for MIMO illustratedin FIG. 5. The following description focuses on these differencesbetween the first to fourth blocks 14000 to 14300 shown in FIG. 14 andthe first to fourth blocks 5000 to 5300 illustrated in FIG. 5.

The constellation mapper block 14010 shown in FIG. 14 can map an inputbit word to a complex symbol. However, the constellation mapper block14010 may not perform constellation rotation, differently from theconstellation mapper block shown in FIG. 5. The constellation mapperblock 14010 shown in FIG. 14 is commonly applicable to the first, secondand third blocks 14000, 14100 and 14200, as described above.

The rotation & I/Q interleaver block 14020 can independently interleavein-phase and quadrature-phase components of each complex symbol ofcell-interleaved data output from the cell interleaver and output thein-phase and quadrature-phase components on a symbol-by-symbol basis.The number of number of input data pieces and output data pieces of therotation & I/Q interleaver block 14020 is two or more which can bechanged by the designer. In addition, the rotation & I/Q interleaverblock 14020 may not interleave the in-phase component.

The rotation & I/Q interleaver block 14020 is commonly applicable to thefirst to fourth blocks 14000 to 14300, as described above. In this case,whether or not the rotation & I/Q interleaver block 14020 is applied tothe fourth block 14300 for processing the PLS-pre/post information canbe signaled through the above-described preamble.

The third block 14200 for MIMO can include a Q-block interleaver block14210 and a complex symbol generator block 14220, as illustrated in FIG.14.

The Q-block interleaver block 14210 can permute a parity part of anFEC-encoded FEC block received from the FEC encoder. Accordingly, aparity part of an LDPC H matrix can be made into a cyclic structure likean information part. The Q-block interleaver block 14210 can permute theorder of output bit blocks having Q size of the LDPC H matrix and thenperform row-column block interleaving to generate final bit streams.

The complex symbol generator block 14220 receives the bit streams outputfrom the Q-block interleaver block 14210, maps the bit streams tocomplex symbols and outputs the complex symbols. In this case, thecomplex symbol generator block 14220 can output the complex symbolsthrough at least two paths. This can be modified by the designer.

The above-described blocks may be omitted or replaced by blocks havingsimilar or identical functions according to design.

The coding & modulation module according to another embodiment of thepresent invention, illustrated in FIG. 14, can output data pipes,PLS-pre information and PLS-post information processed for respectivepaths to the frame structure module.

FIG. 15 illustrates a demapping & decoding module according to anotherembodiment of the present invention.

The demapping & decoding module shown in FIG. 15 corresponds to anotherembodiment of the demapping & decoding module illustrated in FIG. 11.The demapping & decoding module shown in FIG. 15 can perform a reverseoperation of the operation of the coding & modulation module illustratedin FIG. 14.

As shown in FIG. 15, the demapping & decoding module according toanother embodiment of the present invention can include a first block15000 for SISO, a second block 11100 for MISO, a third block 15200 forMIMO and a fourth block 14300 for processing the PLS-pre/PLS-postinformation. In addition, the demapping & decoding module can includeblocks for processing data pipes equally or differently according todesign. The first to fourth blocks 15000 to 15300 shown in FIG. 15 aresimilar to the first to fourth blocks 11000 to 11300 illustrated in FIG.11.

However, the first to fourth blocks 15000 to 15300 shown in FIG. 15 aredistinguished from the first to fourth blocks 11000 to 11300 illustratedin FIG. 11 in that an I/Q deinterleaver and derotation block 15010 ispresent between the time interleaver and the cell deinterleaver of thefirst to fourth blocks 15000 to 15300, a constellation mapper 15010included in the first to fourth blocks 15000 to 15300 has a functiondifferent from the first to fourth blocks 11000 to 11300 illustrated inFIG. 11 and the third block 15200 for MIMO has a configuration differentfrom the third block 11200 for MIMO illustrated in FIG. 11. Thefollowing description focuses on these differences between the first tofourth blocks 15000 to 15300 shown in FIG. 15 and the first to fourthblocks 11000 to 11300 illustrated in FIG. 11.

The I/Q deinterleaver & derotation block 15010 can perform a reverseprocess of the process performed by the rotation & I/Q interleaver block14020 illustrated in FIG. 14. That is, the I/Q deinterleaver &derotation block 15010 can deinterleave I and Q componentsI/Q-interleaved and transmitted by the apparatus for transmittingbroadcast signals and derotate complex symbols having the restored I andQ components.

The I/Q deinterleaver & derotation block 15010 is commonly applicable tothe first to fourth blocks 15000 to 15300, as described above. In thiscase, whether or not the I/Q deinterleaver & derotation block 15010 isapplied to the fourth block 15300 for processing the PLS-pre/postinformation can be signaled through the above-described preamble.

The constellation demapper block 15020 can perform a reverse process ofthe process performed by the constellation mapper block 14010illustrated in FIG. 14. That is, the constellation demapper block 15020can demap cell-deinterleaved data without performing derotation.

The third block 15200 for MIMO can include a complex symbol parsingblock 15210 and a Q-block deinterleaver block 15220, as shown in FIG.15.

The complex symbol parsing block 15210 can perform a reverse process ofthe process performed by the complex symbol generator block 14220illustrated in FIG. 14. That is, the complex symbol parsing block 15210can parse complex data symbols and demap the same to bit data. In thiscase, the complex symbol parsing block 15210 can receive complex datasymbols through at least two paths.

The Q-block deinterleaver block 15220 can perform a reverse process ofthe process carried out by the Q-block interleaver block 14210illustrated in FIG. 14. That is, the Q-block deinterleaver block 15220can restore Q size blocks according to row-column deinterleaving,restore the order of permuted blocks to the original order and thenrestore positions of parity bits to original positions according toparity deinterleaving.

The above-described blocks may be omitted or replaced by blocks havingsimilar or identical functions according to design.

As illustrated in FIG. 15, the demapping & decoding module according toanother embodiment of the present invention can output data pipes andPLS information processed for respective paths to the output processor.

As described above, the apparatus and method for transmitting broadcastsignals according to an embodiment of the present invention canmultiplex signals of different broadcast transmission/reception systemswithin the same RF channel and transmit the multiplexed signals and theapparatus and method for receiving broadcast signals according to anembodiment of the present invention can process the signals in responseto the broadcast signal transmission operation. Accordingly, it ispossible to provide a flexible broadcast transmission and receptionsystem.

FIG. 16 is a conceptual diagram illustrating combinations ofinterleavers on the condition that Signal Space Diversity (SSD) is notconsidered.

When SSD is not considered, combinations of the interleavers may bedenoted by four scenarios S1 to S4. Each scenario may include a cellinterleaver, a time interleaver, and/or a block interleaver.

The scope or spirit of the present invention is not limited tocombinations of the above interleavers, and the present invention canprovide a variety of additional combinations achieved by substitution,deletion, and/or addition of the interleavers. Combinations of theadditional interleavers may be determined in consideration of systemthroughput, receiver operation, memory complexity, robustness, etc. Forexample, a new scenario achieved by omitting the cell interleaver fromeach of four scenarios may be additionally proposed. Although theadditional scenario is not shown in the drawing, the additional scenariois within the scope or spirit of the present invention, and theoperations of this additional scenario may be identical to the sum ofoperation of the individual constituent interleavers.

In FIG. 16, a diagonal time interleaver and a block time interleaver maycorrespond to the above-mentioned time interleavers. In addition, apair-wise frequency interleaver may correspond to an interleavercorresponding to the above-mentioned block interleaver. The individualinterleavers may be a legacy cell interleaver, a legacy time interleaverand/or a legacy block interleaver for use in the conventional art, ormay be a new cell interleaver, a new time interleaver and/or a new blockinterleaver for use in the present invention: The four scenariosmentioned above may include a combination of the legacy interleavers andthe new interleavers. The shaded interleavers shown in FIG. 16 maydenote the new interleavers or may denote the legacy interleavers havingother roles or functions.

TABLE 1 Development Interleaving Single-memory Blocks Types Status SeedVariation Deinterleaving Cell Type-A New YES YES Interleaver Type-BConventional NO (2-period) YES Block Time Type-A Conventional ″ YESInterleaver Type-B Conventional ″ YES Diagonal Type-A New ″ YES TimeType-B New ″ YES Interleaver (pair-wise) ″ New YES YES FrequencyInterleaver

Table 1 shows various interleavers for use in the four scenarios.“Types” item define various types of the respective interleavers. Forexample, the cell interleavers may include a Type-A interleaver and/or aType-B interleaver. The block time interleavers may include a Type-Ainterleaver and/or a Type-B interleaver. “Development Status” item maydenote development states of types of the respective interleavers. Forexample, the Type-A cell interleaver may be a new cell interleaver, andthe Type-B cell interleaver may be a conventional cell interleaver.“Interleaving Seed Variation” item may indicate whether the interleavingseed of each interleaver is changeable. “YES” item may indicate that theinterleaving seed of each interleaver is changeable (i.e., YES). “SingleMemory Deinterleaving” item may indicate whether a deinterleavercorresponding to each interleaver provides single memory deinterleaving.“YES” item may indicate single memory deinterleaving.

A Type-B cell interleaver may correspond to a frequency interleaver foruse in the conventional art (T2, NGH). A Type-A block time interleavermay correspond to DVB-T2. A Type-B block time interleaver may correspondto an interleaver for use in DVB-NGH.

TABLE 2 Blocks Types Key Properties Cell Type-A Different interleavingseed is applied for every FEC Interleaver block Possible to use asingle-memory at receiver Type-B even & odd interleaving seeds areapplied to FEC blocks, in turn Possible to use a single-memory atreceiver (pair-wise) ″ Different interleaving seed is applied for everyFrequency OFDM symbol Interleaver Possible to use a single-memory atreceiver

Table 2 shows a Type-A cell interleaver, a Type-B cell interleaver, anda frequency interleaver. As described above, the frequency interleavermay correspond to the above-mentioned block interleaver.

The basic operation of the cell interleaver shown in Table 1 isidentical to those of Table 2. The cell interleaver may performinterleaving of a plurality of cells corresponding to one FEC block, andoutput the interleaving result. In this case, cells corresponding toindividual FEC blocks may be output in different orders of theindividual FEC blocks. The cell deinterleaver may perform deinterleavingfrom the locations of cells interleaved in one FEC block to the originallocations of the cells. The cell interleaver and the cell deinterleavermay be omitted as described above, or may be replaced with otherblocks/modules having the same or similar functions.

The Type-A cell interleaver is newly proposed by the present invention,and may perform interleaving by applying different interleaving seeds toindividual FEC blocks. Specifically, cells corresponding to one FECblock may be interleaved at intervals of a predetermined time, and theinterleaved resultant cells can be generated. The Type-A celldeinterleaver may perform deinterleaving using a single memory.

The Type-B cell interleaver may be implemented when the interleaver usedas a frequency interleaver for use in the conventional art (T2, NGH) isused as the cell interleaver.

The Type-B cell interleaver may perform interleaving of cellscorresponding to one FEC block, and may output the interleaved cells.The Type-B cell interleaver may apply different interleaving seeds to aneven FEC block and an odd FEC block, and then perform interleaving.Accordingly, the Type-B cell interleaver has a disadvantage in thatdifferent interleaving seeds are applied to individual FEC blocks ascompared to the Type-A cell interleaver. The Type-B cell deinterleavermay perform deinterleaving using a single memory.

A general frequency interleaver may correspond to the above-mentionedblock interleaver. The basic operation of the block interleaver (i.e.,frequency interleaver) is identical to the above-described operations.The block interleaver may perform interleaving of cells contained in atransmission (Tx) block used as a unit of a transmission (Tx) frame soas to obtain an additional diversity gain. The pair-wise blockinterleaver may process two contiguous cells into one unit, and performinterleaving of the processed result. Accordingly, output cells of thepair-wise block interleaver may be two contiguous cells to be arrangedcontiguous to each other. The output cells may operate in the samemanner as in two antenna paths, or may operate independently of eachother.

The operations of a general block deinterleaver (frequencydeinterleaver) may be identical to the basic operations of theabove-mentioned block deinterleaver. The block deinterleaver may performa reverse process of the block interleaver operation so as to recoverthe original data order. The block deinterleaver may performdeinterleaving of data in units of a transmission block (TB). If thepair-wise block interleaver is used by a transmitter, the blockdeinterleaver can perform deinterleaving by pairing two contiguous datapieces of each input path. If deinterleaving is performed by pairing thetwo contiguous data pieces, output data may be two contiguous datapieces. The block interleaver and the block deinterleaver may be omittedas described above, or may be replaced with other blocks/modules havingthe same or similar functions.

The pair-wise frequency interleaver may be a new frequency interleaverproposed by the present invention. The new frequency interleaver mayperform modified operations of the basic operations of theabove-mentioned block interleaver. The new frequency interleaver mayoperate by applying different interleaving seeds to respective OFDMsymbols according to an embodiment. In accordance with anotherembodiment, OFDM symbols are paired so that interleaving may beperformed on the paired OFDM symbols. In this case, differentinterleaving seeds may be applied to one OFDM symbol pair. That is, thesame interleaving seeds may be assigned to the paired OFDM symbols. TheOFDM symbol pair may be implemented by combining two contiguous OFDMsymbols. Two data carriers of the OFDM symbol may be paired andinterleaving may be performed on the paired data carriers.

A new frequency interleaver may perform interleaving using two memories.In this case, the even pair may be interleaved using a first memory, andthe odd pair may be interleaved using a second memory. The pair-wisefrequency deinterleaver may perform deinterleaving using a singlememory. In this case, the pair-wise frequency deinterleaver may indicatea new frequency deinterleaver corresponding to a new frequencyinterleaver.

TABLE 3 Blocks Types Key Properties Block Time Type-A Column-wisewriting and row-wise reading operations Interleaver Actual interleavingdepth of a single FEC block is more than 2 Possible to use asingle-memory at receiver Type-B Column-wise writing and row-wisereading operations Actual interleaving depth of a single FEC block is 1Possible to use a single-memory at receiver Diagonal Time Type-AColumn-wise writing and diagonal-wise reading operations InterleaverActual interleaving depth of a single FEC block is more than 2 Possibleto use a single-memory at receiver Type-B Column-wise writing anddiagonal-wise reading operations Actual interleaving depth of a singleFEC block is 1 Possible to use a single-memory at receiver

Table 3 shows a Type-A block time interleaver, a Type-B block timeinterleaver, a Type-A diagonal time interleaver, and a Type-B diagonaltime interleaver. The diagonal time interleaver and the block timeinterleaver may correspond to the above-mentioned time interleavers.

A general time interleaver may mix the cells corresponding to aplurality of FEC blocks, and output the mixed cells. Cells contained ineach FEC block are scattered by a time interleaving depth through timeinterleaving, and the scattered cells can be transmitted. A diversitygain can be obtained through time interleaving. A general timedeinterleaver may perform a reverse process of the time interleaveroperation. The time deinterleaver may perform deinterleaving of cellsinterleaved in the time domain into the original locations of the cells.The time interleaver and the time deinterleaver may be omitted asdescribed above, or may be replaced with other blocks/modules having thesame or similar functions.

The block time interleaver shown in Table 3 may perform the operationssimilar to those of the time interleaver used in the conventional art(T2, NGH). The Type-A block time interleaver may indicate two or moreinterleavers, each of which has an interleaving depth with respect toone input FEC block. In addition, the type-B block time interleaver mayindicate a specific interleaver which has an interleaving depth of 1with respect to one input FEC block. In this case, the interleavingdepth may indicate a column-wise writing period.

The diagonal time interleaver shown in Table 3 may be a new timeinterleaver proposed by the present invention. The diagonal timeinterleaver may perform the reading operation in a diagonal direction ina different way from the above-mentioned block time interleaver. Thatis, the diagonal time interleaver may store the FEC block in a memory byperforming the column-wise writing operation, and may read the cellsstored in the memory by performing the diagonal-wise reading operation.The number of memories used in the above-mentioned case may be set to 2according to the present invention. The diagonal-wise reading operationmay indicate the operation for reading the cells diagonally spaced apartfrom each other by a predetermined distance in the interleaving arraystored in the memory. Interleaving may be achieved through thediagonal-wise reading operation. The diagonal time interleaver may becalled a twisted row-column block interleaver.

The Type-A diagonal time interleaver may indicate an interleaver havingan interleaving depth of 2 or higher with respect to one input FECblock. In addition, the Type-B diagonal time interleaver may indicate aninterleaver having an interleaving depth of 1 with respect to one inputFEC block. In this case, the interleaving depth may indicate thecolumn-wise writing period.

FIG. 17 shows the column-wise writing operations of the block timeinterleaver and the diagonal time interleaver according to the presentinvention.

The column-wise writing operation of the Type-A block time interleaverand the Type-A diagonal time interleaver may have the interleaving depthof 2 or higher as shown in FIG. 17.

The column-wise writing operation of the Type-B block time interleaverand the Type-B diagonal time interleaver may have the interleaving depthof 1 as shown in FIG. 17. In this case, the interleaving depth mayindicate the column-wise writing period.

FIG. 18 is a conceptual diagram illustrating a first scenario S2 fromamong combinations of the interleavers without consideration of a signalspace diversity (SSD).

FIG. 18(a) shows the interleaving structure according to the firstscenario. The interleaving structure of the first scenario may include aType-B cell interleaver, a Type-A or Type-B diagonal time interleaver,and/or a pair-wise frequency interleaver. In this case, the pair-wisefrequency interleaver may be the above-mentioned new frequencyinterleaver.

The Type-B cell interleaver may mix the cells corresponding to one FECblock at random, and output the mixed cells. In this case, the cellscorresponding to each FEC block may be output in different orders ofindividual FEC blocks. The Type-B cell interleaver may performinterleaving by applying different interleaving seeds to odd input FECblocks and even input FEC blocks as described above. The cellinterleaving can be implemented by performing not only the writingoperation for writing data in the memory, but also the reading operationfor reading data from the memory.

The Type-A and Type-B diagonal time interleavers may perform thecolumn-wise writing operation and the diagonal-wise reading operationfor the cells belonging to a plurality of FEC blocks. Cells located atother locations within each FEC block through the diagonal timeinterleaving are scattered and transmitted within an interval as long asa diagonal interleaving depth, such that a diversity gain can beobtained.

Thereafter, the output of the diagonal time interleaver may be input tothe pair-wise frequency interleaver after passing through otherblocks/modules such as the above-mentioned cell mapper or the like. Inthis case, the pair-wise frequency interleaver may be a new frequencyinterleaver. Accordingly, the pair-wise frequency interleaver (newfrequency interleaver) may provide an additional diversity gain byinterleaving the cells contained in the OFDM symbol.

FIG. 18(b) shows the deinterleaving structure according to the firstscenario. The deinterleaving structure of the first scenario may includea (pair-wise) frequency deinterleaver, a Type-A or Type-B diagonal timedeinterleaver, and/or a Type-B cell deinterleaver. In this case, thepair-wise frequency deinterleaver may correspond to the above-mentionednew frequency deinterleaver. The pair-wise frequency deinterleaver mayperform deinterleaving of data through a reverse process of the newfrequency interleaver operation.

Thereafter, the output of the pair-wise frequency deinterleaver may beinput to the Type-A and Type-B diagonal time deinterleavers afterpassing through other blocks/modules such as the above-mentioned celldemapper. The Type-A diagonal time deinterleaver may perform a reverseprocess of the Type-A diagonal time interleaver. The Type-B diagonaltime deinterleaver may perform a reverse process of the Type-B diagonaltime interleaver. In this case, the Type-A and Type-B diagonal timedeinterleaver may perform time deinterleaving using a single memory.

The Type-B cell deinterleaver may perform deinterleaving from thelocations of the cells interleaved in one FEC block to the originallocations of the cells.

FIG. 19 is a conceptual diagram of a second scenario S2 from amongcombinations of the interleavers without consideration of a signal spacediversity (SSD).

FIG. 19(a) shows the interleaving structure according to the secondscenario. The interleaving structure of the second scenario may includea Type-A cell interleaver, a Type-A or Type-B block time interleaver,and/or a pair-wise frequency interleaver. In this case, the pair-wisefrequency interleaver may be the above-mentioned new frequencyinterleaver.

The Type-A cell interleaver may perform interleaving by applyingdifferent interleaving seeds to respective input FEC blocks as describedabove.

The Type-A and Type-B block timer interleavers may perform interleavingof the cells belonging to a plurality of FEC blocks through thecolumn-wise writing operation and the row-wise reading operation, asdescribed above. Cells located at other locations within are scatteredand transmitted within an interval as long as an interleaving depth,such that a diversity gain can be obtained.

Thereafter, the output of the block time interleaver may be input to thepair-wise frequency interleaver after passing through otherblocks/modules such as the above-mentioned cell mapper or the like. Inthis case, the pair-wise frequency interleaver may be theabove-mentioned new frequency interleaver. Accordingly, the pair-wisefrequency interleaver (new frequency interleaver) may provide anadditional diversity gain by interleaving the cells contained in theOFDM symbol.

FIG. 19(b) shows the deinterleaving structure according to the secondscenario. The deinterleaving structure of the second scenario mayinclude a (pair-wise) frequency deinterleaver, a Type-A or Type-Bdiagonal time deinterleaver, and/or a Type-A cell deinterleaver. In thiscase, the pair-wise frequency deinterleaver may correspond to theabove-mentioned new frequency deinterleaver.

The pair-wise frequency deinterleaver may perform deinterleaving of datathrough a reverse process of the new frequency interleaver operation.

Thereafter, the output of the pair-wise frequency deinterleaver may beinput to the Type-A and Type-B diagonal time deinterleavers afterpassing through other blocks/modules such as the above-mentioned celldemapper. The Type-A block time deinterleaver may perform a reverseprocess of the Type-A block time interleaver. The Type-B block timedeinterleaver may perform a reverse process of the Type-B block timeinterleaver. In this case, the Type-A or Type-B block time deinterleavermay perform time deinterleaving using a single memory.

The Type-A cell deinterleaver may perform deinterleaving from thelocations of the cells interleaved in one FEC block to the originallocations of the cells.

FIG. 20 is a conceptual diagram of a third scenario S3 from amongcombinations of the interleavers without consideration of signal spacediversity (SSD).

FIG. 20(a) shows the interleaving structure according to the thirdscenario. The interleaving structure of the third scenario may include aType-A cell interleaver, a Type-A or Type-B diagonal time interleaver,and/or a pair-wise frequency interleaver. In this case, the pair-wisefrequency interleaver may be the above-mentioned new frequencyinterleaver.

The operations of the Type-A cell interleaver, the Type-A and Type-Bdiagonal time interleaver, and the pair-wise frequency interleaver maybe identical to those of the above-mentioned figures.

FIG. 19(b) shows the deinterleaving structure according to the thirdscenario. The deinterleaving structure of the third scenario may includea (pair-wise) frequency deinterleaver, a Type-A or Type-B diagonal timedeinterleaver, and/or a Type-A cell deinterleaver. In this case, thepair-wise frequency deinterleaver may correspond to the above-mentionednew frequency deinterleaver.

The operations of the pair-wise frequency deinterleaver, the Type-A andType-B diagonal time interleavers, and the Type-A cell deinterleaver maybe identical to those of the above-mentioned figures.

FIG. 21 is a conceptual diagram of a fourth scenario S4 from amongcombinations of the interleavers without consideration of a signal spacediversity (SSD).

FIG. 21(a) shows the interleaving structure according to the fourthscenario. The interleaving structure of the fourth scenario may includea Type-A or Type-B diagonal time interleaver and/or a pair-wisefrequency interleaver. In this case, the pair-wise frequency interleavermay be the above-mentioned new frequency interleaver.

The operations of the Type-A and Type-B diagonal time interleavers andthe pair-wise frequency deinterleaver may be identical to those of theabove-mentioned figures.

FIG. 21(b) shows the deinterleaving structure according to the fourthscenario. The deinterleaving structure of the fourth scenario mayinclude a (pair-wise) frequency deinterleaver and/or a Type-A or Type-Bdiagonal time deinterleaver. In this case, the pair-wise frequencydeinterleaver may correspond to the above-mentioned new frequencydeinterleaver.

The operations of the pair-wise frequency deinterleaver and the Type-Aor Type-B diagonal time interleaver may be identical to those of theabove-mentioned figures.

FIG. 22 illustrates a structure of a random generator according to anembodiment of the present invention.

FIG. 22 illustrates the case in which the random generator generates aninitial-offset value using a PP method.

The random generator according to an embodiment of the present inventionmay include a register 32000 and an XOR operator 32100. In general, thePP method may randomly output values 1, . . . , 2n−1. Accordingly, therandom generator according to an embodiment of the present invention mayperform a register reset process in order to output 2^(n) symbol indexesincluding 0 and set a register initial value for a register shiftingprocess.

The random generator according to an embodiment of the present inventionmay include different registers and XOR operators for respectiveprimitive polynomials for the PP method.

Table 4 below shows primitive polynomials for the aforementioned PPmethod and a reset value and an initial value for the register resetprocess and the register shifting process.

TABLE 4 Order (n) Primitive polynomial k = 0 (reset value) k = 1(initial value) 9 f(x) = 1 + x⁵ + x⁹ [0 0 0 0 0 0 0 0 0] [0 0 0 0 1 0 00 1] 10 f(x) = 1 + x⁷ + x¹⁰ [0 0 0 0 0 0 0 0 0 0] [0 0 0 0 0 0 1 0 0 1]11 f(x) = 1 + x⁹ + x¹¹ [0 0 0 0 0 0 0 0 0 0 0] [0 0 0 0 0 0 0 0 1 0 1]12 f(x) = 1 + x⁶ + x⁸ + x¹¹ + x¹² [0 0 0 0 0 0 0 0 0 0 0 0] [0 0 0 0 0 10 1 0 0 1 1] 13 f(x) = 1 + x² + x⁴ + x⁸ + x⁹ + x¹² + x¹³ [0 0 0 0 0 0 00 0 0 0 0 0] [0 1 0 1 0 0 0 1 1 0 0 1 1] 14 f(x) = 1 + x² + x¹² + x¹³ +x¹⁴ [0 0 0 0 0 0 0 0 0 0 0 0 0 0] [0 0 1 0 0 0 0 0 0 0 0 1 1 1] 15 f(x)= 1 + x¹⁴ + x¹⁵ [0 0 0 0 0 0 0 0 0 0 0 0 0 0 0] [0 0 0 0 0 0 0 0 0 0 0 00 1 1]

Table 4 above shows a register reset value and register initial valuecorresponding to an n^(th) primitive polynomial (n=9, . . . , 15). Asshown in Table 4 above, k=0 refers to a register reset value and k=1refers to a register initial value. In addition, 2≦k≦2^(n)−1 refers toshifted register values.

FIG. 23 illustrates a random generator according to an embodiment of thepresent invention.

FIG. 23 illustrates a structure of the random generator when n of then^(th) primitive polynomial of Table 4 above is 9 to 12.

FIG. 24 illustrates a random generator according to another embodimentof the present invention.

FIG. 24 illustrates a structure of the random generator when n of then^(th) primitive polynomial of Table 4 above is 13 to 15.

FIG. 25 illustrates a frequency interleaving process according to anembodiment of the present invention.

FIG. 25 illustrates a frequency interleaving process when a singlememory is applied to a broadcast signal receiver, if the number of allsymbols is 10, the number of cells included in one symbol is 10, and pis 3, according to an embodiment of the present invention.

FIG. 25(a) illustrates output values of respective symbols, which isoutput using an RPI method. In particular, a first memory index value ofeach OFDM symbol, that is, 0, 7, 4, 1, 8 . . . may be set as aninitial-offset value generated by the random generator of theaforementioned RPI. A number indicated in the interleaving memory indexrepresents an order in which cells included in each symbol areinterleaved and output.

FIG. 25(b) illustrates results obtained by interleaving cells of aninput OFDM symbol in a symbol unit using the generated interleavingmemory index.

FIG. 26 is a conceptual diagram illustrating a frequency deinterleavingprocess according to an embodiment of the present invention.

FIG. 26 illustrates a frequency deinterleaving process when a singlememory is applied to a broadcast signal receiver and, that is, anembodiment in which the number of cells included in one symbol is 10.

The broadcast signal receiver (or a frame parsing module or a blockinterleaver) according to an embodiment of the present invention maygenerate a deinterleaving memory index via a process of sequentiallywriting symbols interleaved via the aforementioned frequencyinterleaving in an input order and output deinterleaved symbols via areading process. In this case, the broadcast signal receiver accordingto an embodiment of the present invention may perform a process ofperforming writing on a deinterleaving memory index on which the readingis performed.

FIG. 27 illustrates a frequency deinterleaving process according to anembodiment of the present invention.

FIG. 27 illustrates a deinterleaving process when the number of allsymbols is 10, the number of cells included in one symbol is 10, and pis 3.

FIG. 27(a) illustrates symbols input to a single memory according to anembodiment of the present invention. That is, the single-memory inputsymbols shown in FIG. 27(a) refer to values stored in the single-memoryaccording to each input symbol. In this case, the values stored in thesingle-memory according to each input symbol refer to a result obtainedby sequentially writing currently input symbol cells while reading aprevious symbol.

FIG. 27(b) illustrates a process of generation a deinterleaving memoryindex.

The deinterleaving memory index is an index used to deinterleave valuesstored in a single memory, and a number indicated in the deinterleavingmemory index refers to an order in which cells included in each symbolare deinterleaved and output.

Hereinafter, the aforementioned frequency deinterleaving process will bedescribed in terms of input symbols #0 and #1 among illustrated symbols.

The broadcast signal receiver according to an embodiment of the presentinvention sequentially writes input symbol #0 in a single memory. Thenthe broadcast signal receiver according to an embodiment of the presentinvention may sequentially generate the aforementioned deinterleavingmemory index in an order of 0, 3, 6, 9 . . . in order to deinterleaveinput symbol #0.

Then the broadcast signal receiver according to an embodiment of thepresent invention reads input symbol #0 written (or stored) in thesingle memory according to the generated deinterleaving memory index.The already written values do not have to be stored and thus a newlyinput symbol #1 may be sequentially re-written.

Then the process of reading input symbol #1 and the process of writinginput symbol #1 are completed, the deinterleaving memory index may begenerated in order to deinterleave the written input symbol #1. In thiscase, since the broadcast signal receiver according to an embodiment ofthe present invention uses a single memory, interleaving cannot beperformed using an interleaving pattern applied to each symbol appliedin the broadcast signal transmitter. Then deinterleaving processing canbe performed on input symbols in the same way.

FIG. 28 illustrates a process of generating a deinterleaved memory indexaccording to an embodiment of the present invention.

In particular, FIG. 28 illustrates a method of generating a newinterleaving pattern when interleaving cannot be performed using aninterleaving pattern applied to each symbol applied in the broadcastsignal transmitter since the broadcast signal receiver according to anembodiment of the present invention users a single memory.

FIG. 28(a) illustrates a deinterleaving memory index of a j^(th) inputsymbol and FIG. 28(b) illustrates the aforementioned process ofgenerating a deinterleaving memory index together with Math Figures.

As shown in FIG. 28(b), according to an embodiment of the presentinvention, a variable of RPI of each input symbol is used.

According to an embodiment of the present invention, a process ofgenerating a deinterleaving memory index of input symbol #0 uses p=3 andI₀=0 as a variable of RPI like in the broadcast signal transmitter.According to an embodiment of the present invention, in the case ofinput symbol #1, p²=3×3 and I₀=1 may be used as a variable of RPI, andin the case of input symbol #2, p³=3×3×3 and I₀=7 may be used as avariable of RPI. In addition, according to an embodiment of the presentinvention, in the case of input symbol #3, p⁴=3×3×3×3 and I₀=4 may beused as a variable of RPI.

That is, the broadcast signal receiver according to an embodiment of thepresent invention may change a value p of RPI and an initial offsetvalue for each symbol and may effectively perform deinterleaving inorder to deinterleave symbols stored in each single memory. In addition,a value p used in each symbol may be easily induced using exponentiationof p and initial offset values may be sequentially acquired using amother interleaving seed. Hereinafter, a method of calculating aninitial offset value will be described.

According to an embodiment of the present invention, an initial offsetvalue used in input symbol #0 is defined as I₀=0. An initial offsetvalue used in input symbol #1 is I₀=1 that is the same as a seventhvalue generated in the deinterleaving memory index generation process ofinput symbol #0. That is, the broadcast signal receiver according to anembodiment of the present invention may store and use the value in thedeinterleaving memory index generation process of input symbol #0.

An initial offset value used in input symbol #2 is I₀=7 that is the sameas a fourth value generated in the deinterleaving memory indexgeneration process of input symbol #1, and an initial offset value usedin input symbol #3 is I₀=4 that is the same as a first value generatedin the deinterleaving memory index generation process of input symbol#2.

Accordingly, the broadcast signal receiver according to an embodiment ofthe present invention may store and use a value corresponding to aninitial offset value to be used in each symbol in a process ofgenerating a deinterleaving memory index of a previous symbol.

As a result, the aforementioned method may be represented according toMath FIG. 1 below.

π_(j) ⁻¹(k)=(I ₁ ⁻¹ +p ^(j+1) k)mod N _(Cell) _(_) _(NUM), for k=0, . .. , N _(Cell) _(_) _(NUM)−1,j=0, . . . ,N _(Sym) _(_) _(NUM)−1  [MathFIG. 1]

where I_(i) ⁻¹=π_(j−1) ⁻¹ (ω(j)) with I₀ ⁻¹=0

I_(j) ⁻¹: the initial-offset value at the j^(th) RPI for deinterleaving

π_(j) ⁻¹(k): deinterleaving output memory-index for the k^(th) inputcell-index in the j^(th) OFDM symbol

π_(j) ⁻¹(ω)): the ω(j)th deinterleaving output memory-index in thej^(th) OFDM symbol

In this case, a position of a value corresponding to each initial offsetvalue may be easily induced according to Math FIG. 1 above.

According to an embodiment of the present invention, the broadcastsignal transmitter according to an embodiment of the present inventionmay recognize two adjacent cells as one cell and perform frequencyinterleaving. This may be referred to as pair-wise interleaving. In thiscase, since two adjacent cells are considered as one cell andinterleaving is performed, it is advantageous that a number of times ofgenerating a memory index may be reduced in half.

Math FIG. 2 below represents the pair-wise RPI.

π_(j)(k)=(ω(j)+pk)mod(N _(Cell) _(_) _(NUM)/2) for k=0, . . . ,N _(Cell)_(_) _(NUM)/2−1,j=0, . . . ,N _(Sym) _(_) _(NUM)−1   [Math FIG. 2]

Math FIG. 3 below represents a pair-wise deinterleaving method.

π_(j) ⁻¹(k)=(I _(j) ⁻¹ +p ^(j+1) k)mod(N _(Cell) _(_) _(NUM)/2) for k=0,. . . ,N _(Cell) _(_) _(NUM)/2−1,j=0, . . . ,N _(Sym) _(_)_(NUM)−1  [Math FIG. 3]

where I_(i) ⁻¹=π_(j−1) ⁻¹(ω(j)) with I₀ ⁻¹=0

FIG. 29 illustrates a frequency interleaving process according to anembodiment of the present invention.

FIG. 29 illustrates an interleaving method for improving frequencydiversity performance using different relative primes including aplurality of OFDM symbols by the aforementioned frequency interleaver.

That is, as shown in FIG. 29, a relative prime value is changed everyframe/super frame so as to further improve a frequency diversityperformance, especially avoiding a repeated interleaving pattern.

The apparatus for receiving broadcast signals according to an embodimentof the present invention can output process the decoded DP data. Morespecifically, the apparatus for receiving broadcast signals according toan embodiment of the present invention can decompress a header in theeach of the data packets in the decoded DP data according to a headercompression mode and recombine the data packets. Details are asdescribed in FIGS. 16 to 32.

FIG. 30 illustrates a super-frame structure according to an embodimentof the present invention.

The apparatus for transmitting broadcast signals according to anembodiment of the present invention can sequentially transmit aplurality of super-frames carrying data corresponding to a plurality ofbroadcast services.

As shown in FIG. 30, frames 17100 of different types and a futureextension frame (FEF) 17110 can be multiplexed in the time domain andtransmitted in a super-frame 17000. The apparatus for transmittingbroadcast signals according to an embodiment of the present inventioncan multiplex signals of different broadcast services on aframe-by-frame basis and transmit the multiplexed signals in the same RFchannel, as described above. The different broadcast services mayrequire different reception conditions or different coverages accordingto characteristics and purposes thereof. Accordingly, signal frames canbe classified into types for transmitting data of different broadcastservices and data included in the signal frames can be processed bydifferent transmission parameters. In addition, the signal frames canhave different FFT sizes and guard intervals according to broadcastservices transmitted through the signal frames. The FEF 17110 shown inFIG. 30 is a frame available for future new broadcast service systems.

The signal frames 17100 of different types according to an embodiment ofthe present invention can be allocated to a super-frame according todesign. Specifically, the signal frames 17100 of different types can berepeatedly allocated to the super-frame in a multiplexed pattern.Otherwise, a plurality of signal frames of the same type can besequentially allocated to a super-frame and then signal frames of adifferent type can be sequentially allocated to the super-frame. Thesignal frame allocation scheme can be changed by the designer.

Each signal frame can include a preamble 17200, an edge data OFDM symbol17210 and a plurality of data OFDM symbols 17220, as shown in FIG. 30.

The preamble 17200 can carry signaling information related to thecorresponding signal frame, for example, a transmission parameter. Thatis, the preamble carries basic PLS data and is located in the beginningof a signal frame. In addition, the preamble 17200 can carry the PLSdata described with reference to FIG. 1. That is, the preamble can carryonly basic PLS data or both basic PLS data and the PLS data describedwith reference to FIG. 1. The information carried through the preamblecan be changed by the designer. The signaling information carriedthrough the preamble can be referred to as preamble signalinginformation.

The edge data OFDM symbol 17210 is an OFDM symbol located at thebeginning or end of the corresponding frame and can be used to transmitpilots in all pilot carriers of data symbols. The edge data OFDM symbolmay be in the form of a known data sequence or a pilot. The position ofthe edge data OFDM symbol 17210 can be changed by the designer.

The plurality of data OFDM symbols 17220 can carry data of broadcastservices.

Since the preamble 17200 illustrated in FIG. 30 includes informationindicating the start of each signal frame, the apparatus for receivingbroadcast signals according to an embodiment of the present inventioncan detect the preamble 17200 to perform synchronization of thecorresponding signal frame. Furthermore, the preamble 17200 can includeinformation for frequency synchronization and basic transmissionparameters for decoding the corresponding signal frame.

Accordingly, even if the apparatus for receiving broadcast signalsaccording to an embodiment of the present invention receives signalframes of different types multiplexed in a super-frame, the apparatusfor receiving broadcast signals can discriminate signal frames bydecoding preambles of the signal frames and acquire a desired broadcastservice.

That is, the apparatus for receiving broadcast signals according to anembodiment of the present invention can detect the preamble 17200 in thetime domain to check whether or not the corresponding signal is presentin the broadcast signal transmission and reception system according toan embodiment of the present invention. Then, the apparatus forreceiving broadcast signals according to an embodiment of the presentinvention can acquire information for signal frame synchronization fromthe preamble 17200 and compensate for a frequency offset. Furthermore,the apparatus for receiving broadcast signals according to an embodimentof the present invention can decode signaling information carried by thepreamble 17200 to acquire basic transmission parameters for decoding thecorresponding signal frame. Then, the apparatus for receiving broadcastsignals according to an embodiment of the present invention can obtaindesired broadcast service data by decoding signaling information foracquiring broadcast service data transmitted through the correspondingsignal frame.

FIG. 31 illustrates a preamble insertion block according to anembodiment of the present invention.

The preamble insertion block illustrated in FIG. 31 corresponds to anembodiment of the preamble insertion block 7500 described with referenceto FIG. 7 and can generate the preamble described in FIG. 30.

As shown in FIG. 31, the preamble insertion block according to anembodiment of the present invention can include a signaling sequenceselection block 18000, a signaling sequence interleaving block 18100, amapping block 18200, a scrambling block 18300, a carrier allocationblock 18400, a carrier allocation table block 18500, an IFFT block18600, a guard insertion block 18700 and a multiplexing block 18800.Each block may be modified or may not be included in the preambleinsertion block by the designer. A description will be given of eachblock of the preamble insertion block.

The signaling sequence selection block 18000 can receive the signalinginformation to be transmitted through the preamble and select asignaling sequence suitable for the signaling information.

The signaling sequence interleaving block 18100 can interleave signalingsequences for transmitting the input signaling information according tothe signaling sequence selected by the signaling sequence selectionblock 18000. Details will be described later.

The mapping block 18200 can map the interleaved signaling informationusing a modulation scheme.

The scrambling block 18300 can multiply mapped data by a scramblingsequence.

The carrier allocation block 18400 can allocate the data output from thescrambling block 18300 to predetermined carrier positions using activecarrier position information output from the carrier allocation tableblock 18500.

The IFFT block 18600 can transform the data allocated to carriers,output from the carrier allocation block 18400, into an OFDM signal inthe time domain.

The guard insertion block 18700 can insert a guard interval into theOFDM signal.

The multiplexing block 18800 can multiplex the signal output from theguard insertion block 18700 and a signal c(t) output from the guardsequence insertion block 7400 illustrated in FIG. 7 and output an outputsignal p(t). The output signal p(t) can be input to the waveformprocessing block 7600 illustrated in FIG. 7.

FIG. 32 illustrates a preamble structure according to an embodiment ofthe present invention.

The preamble shown in FIG. 32 can be generated by the preamble insertionblock illustrated in FIG. 31.

The preamble according to an embodiment of the present invention has astructure of a preamble signal in the time domain and can include ascrambled cyclic prefix part 19000 and an OFDM symbol 19100. Inaddition, the preamble according to an embodiment of the presentinvention may include an OFDM symbol and a scrambled cyclic postfixpart. In this case, the scrambled cyclic postfix part may follow theOFDM symbol, differently from the scrambled prefix, and may be generatedthrough the same process as the process for generating the scrambledcyclic prefix, which will be described later. The position andgeneration process of the scrambled cyclic postfix part may be changedaccording to design.

The scrambled cyclic prefix part 19000 shown in FIG. 32 can be generatedby scrambling part of the OFDM symbol or the whole OFDM symbol and canbe used as a guard interval.

Accordingly, the apparatus for receiving broadcast signals according toan embodiment of the present invention can detect a preamble throughguard interval correlation using a guard interval in the form of acyclic prefix even when a frequency offset is present in a receivedbroadcast signal since frequency synchronization cannot be performed.

In addition, the guard interval in the scrambled cyclic prefix formaccording to an embodiment of the present invention can be generated bymultiplying (or combining) the OFDM symbol by a scrambling sequence (orsequence). Or the guard interval in the scrambled cyclic prefix formaccording to an embodiment of the present invention can be generated byscrambling the OFDM symbol with a scrambling sequence (or sequence). Thescrambling sequence according to an embodiment of the present inventioncan be a signal of any type which can be changed by the designer.

The method of generating the guard interval in the scrambled cyclicprefix form according to an embodiment of the present invention has thefollowing advantages.

Firstly, a preamble can be easily detected by discriminating the guardinterval from a normal OFDM symbol. As described above, the guardinterval in the scrambled cyclic prefix form is generated by beingscrambled by the scrambling sequence, distinguished from the normal OFDMsymbol. In this case, if the apparatus for receiving broadcast signalsaccording to an embodiment of the present invention performs guardinterval correlation, the preamble can be easily detected since only acorrelation peak according to the preamble is generated without acorrelation peak according to the normal OFDM symbol.

Secondly, when the guard interval in the scrambled cyclic prefix formaccording to an embodiment of the present invention is used, a dangerousdelay problem can be solved. For example, if the apparatus for receivingbroadcast signals performs guard interval correlation when multi-pathinterference delayed by the duration Tu of the OFDM symbol is present,preamble detection performance may be deteriorated since a correlationvalue according to multiple paths is present at all times. However, whenthe apparatus for receiving broadcast signals according to an embodimentof the present invention performs guard interval correlation, theapparatus for receiving broadcast signals can detect the preamblewithout being affected by the correlation value according to multiplepaths since only a peak according to the scrambled cyclic prefix isgenerated, as described above.

Finally, the influence of continuous wave (CW) interference can beprevented. If a received signal includes CW interference, the signaldetection performance and synchronization performance of the apparatusfor receiving broadcast signals can be deteriorated since a DC componentcaused by CW is present at all times when the apparatus for receivingbroadcast signals performs guard interval correlation. However, when theguard interval in the scrambled cyclic prefix form according to anembodiment of the present invention is used, the influence of CW can beprevented since the DC component caused by CW is averaged out by thescrambling sequence.

FIG. 33 illustrates a preamble detector according to an embodiment ofthe present invention.

The preamble detector shown in FIG. 33 corresponds to an embodiment ofthe preamble detector 9300 included in the synchronization &demodulation module illustrated in FIG. 9 and can detect the preambleillustrated in FIG. 30.

As shown in FIG. 33, the preamble detector according to an embodiment ofthe present invention can include a correlation detector 20000, an FFTblock 20100, an ICFO (integer carrier frequency offset) estimator 20200,a carrier allocation table block 20300, a data extractor 20300 and asignaling decoder 20500. Each block may be modified or may not beincluded in the preamble detector according to design. A descriptionwill be given of operation of each block of the preamble detector.

The correlation detector 20000 can detect the above-described preambleand estimate frame synchronization, OFDM symbol synchronization, timinginformation and FCFO (fractional frequency offset). Details will bedescribed later.

The FFT block 20100 can transform the OFDM symbol part included in thepreamble into a frequency domain signal using the timing informationoutput from the correlation detector 20000.

The ICFO estimator 20200 can receive position information on activecarriers, output from the carrier allocation table block 20300, andestimate ICFO information.

The data extractor 20300 can receive the ICFO information output fromthe ICFO estimator 20200 to extract signaling information allocated tothe active carriers and the signaling decoder 20500 can decode theextracted signaling information.

Accordingly, the apparatus for receiving broadcast signals according toan embodiment of the present invention can obtain the signalinginformation carried by the preamble through the above-describedprocedure.

FIG. 34 illustrates a correlation detector according to an embodiment ofthe present invention.

The correlation detector shown in FIG. 34 corresponds to an embodimentof the correlation detector illustrated in FIG. 33.

The correlation detector according to an embodiment of the presentinvention can include a delay block 21000, a conjugate block 21100, amultiplier, a correlator block 21200, a peak search block 21300 and anFCFO estimator block 21400. A description will be given of operation ofeach block of the correlation detector.

The delay block 21000 of the correlation detector can delay an inputsignal r(t) by the duration Tu of the OFDM symbol in the preamble.

The conjugate block 21100 can perform conjugation on the delayed signalr(t).

The multiplier can multiply the signal r(t) by the conjugated signalr(t) to generate a signal m(t).

The correlator block 21200 can correlate the signal m(t) input theretoand the scrambling sequence to generate a descrambled signal c(t).

The peak search block 21300 can detect a peak of the signal c(t) outputfrom the correlator block 21200. In this case, since the scrambledcyclic prefix included in the preamble is descrambled by the scramblingsequence, a peak of the scrambled cyclic prefix can be generated.However, OFDM symbols or components caused by multiple paths other thanthe scrambled cyclic prefix are scrambled by the scrambling sequence,and thus a peak of the OFDM symbols or components caused by multiplepaths is not generated. Accordingly, the peak search block 21300 caneasily detect the peak of the signal c(t).

The FCFO estimator block 21400 can acquire frame synchronization andOFDM symbol synchronization of the signal input thereto and estimateFCFO information from a correlation value corresponding to the peak.

As described above, the scrambling sequence according to an embodimentof the present invention can be a signal of any type and can be changedby the designer.

FIGS. 21 to 25 illustrate results obtained when a chirp-like sequence, abalanced m-sequence, a Zadoff-Chu sequence and a binary chirp-likesequence are used as the scrambling sequence according to an embodimentof the present invention.

Each figure will now be described.

FIG. 35 shows graphs representing results obtained when the scramblingsequence according to an embodiment of the present invention is used.

The graph of FIG. 35 shows results obtained when the scrambling sequenceaccording to an embodiment of the present invention is a chirp-likesequence. The chirp-like sequence can be calculated according to MathFIG. 4.

e ^(j2πk/80) for k=0˜79,

e ^(j2πk/144) for k=80˜223,

e ^(j2πk/272) for k=224˜495,

e ^(j2πk/528) for k=496˜1023  [Math FIG. 4]

As represented by Math FIG. 4, the chirp-like sequence can be generatedby connecting sinusoids of 4 different frequencies corresponding to oneperiod.

As shown in FIG. 35, (a) is a graph showing waveforms of the chirp-likesequence according to an embodiment of the present invention.

The first waveform 22000 shown in (a) represents a real number part ofthe chirp-like sequence and the second waveform 22100 represents animaginary number part of the chirp-like sequence. The duration of thechirp-like sequence corresponds to 1024 samples and the averages of areal number part sequence and an imaginary number part sequence are 0.

As shown in FIG. 35, (b) is a graph showing the waveform of the signalc(t) output from the correlator block illustrated in FIGS. 20 and 21when the chirp-like sequence is used.

Since the chirp-like sequence is composed of signals having differentperiods, dangerous delay is not generated. Furthermore, the correlationproperty of the chirp-like sequence is similar to guard intervalcorrelation and thus distinctly discriminated from the preamble ofconventional broadcast signal transmission/reception systems.Accordingly, the apparatus for receiving broadcast signals according toan embodiment of the present invention can easily detect the preamble.In addition, the chirp-like sequence can provide correct symbol timinginformation and is robust to noise on a multi-path channel, compared toa sequence having a delta-like correlation property, such as anm-sequence. Furthermore, when scrambling is performed using thechirp-like sequence, it is possible to generate a signal having abandwidth slightly increased compared to the original signal.

FIG. 36 shows graphs representing results obtained when a scramblingsequence according to another embodiment of the present invention isused.

The graphs of FIG. 36 are obtained when the balanced m-sequence is usedas a scrambling sequence. The balanced m-sequence according to anembodiment of the present invention can be calculated by Math FIG. 5.

g(x)=x ¹⁰ +x ⁸ +X ⁴ +x ³+1  [Math FIG. 5]

The balanced m-sequence can be generated by adding a sample having avalue of ‘+1’ to an m-sequence having a length corresponding to 1023samples according to an embodiment of the present invention. The lengthof balanced m-sequence is 1024 samples and the average thereof is ‘0’according to one embodiment. The length and average of the balancedm-sequence can be changed by the designer.

As shown in FIG. 36, (a) is a graph showing the waveform of the balancedm-sequence according to an embodiment of the present invention and (b)is a graph showing the waveform of the signal c(t) output from thecorrelator block illustrated in FIGS. 20 and 21 when the balancedm-sequence is used.

When the balanced m-sequence according to an embodiment of the presentinvention is used, the apparatus for receiving broadcast signalsaccording to an embodiment of the present invention can easily performsymbol synchronization on a received signal since preamble correlationproperty corresponds to a delta function.

FIG. 37 shows graphs representing results obtained when a scramblingsequence according to another embodiment of the present invention isused.

The graphs of FIG. 37 show results obtained when the Zadoff-Chu sequenceis used as a scrambling sequence. The Zadoff-Chu sequence according toan embodiment of the present invention can be calculated by Math FIG. 6.

e ^(−jπuk(k+1)/1023) for k=0˜1022,u=23  [Math FIG. 6]

The Zadoff-Chu sequence may have a length corresponding to 1023 samplesand u value of 23 according to one embodiment. The length and u value ofthe Zadoff-Chu sequence can be changed by the designer.

As shown in FIG. 37, (a) is a graph showing the waveform of the signalc(t) output from the correlator block illustrated in FIGS. 20 and 21when the Zadoff-Chu sequence according to an embodiment of the presentinvention is used.

As shown in FIG. 37, (b) is a graph showing the in-phase waveform of theZadoff-Chu sequence according to an embodiment of the present inventionand (c) is a graph showing the quadrature phase waveform of theZadoff-Chu sequence according to an embodiment of the present invention.

When the Zadoff-Chu sequence according to an embodiment of the presentinvention is used, the apparatus for receiving broadcast signalsaccording to an embodiment of the present invention can easily performsymbol synchronization on a received signal since preamble correlationproperty corresponds to a delta function. In addition, the envelope ofthe received signal is uniform in both the frequency domain and timedomain.

FIG. 38 is a graph showing a result obtained when a scrambling sequenceaccording to another embodiment of the present invention is used. Thegraph of FIG. 38 shows waveforms of a binary chirp-like sequence. Thebinary chirp-like sequence is an embodiment of the signal that can beused as the scrambling sequence according to the present invention.

$\begin{matrix}{{x\lbrack k\rbrack} = \left\{ {{i\lbrack k\rbrack},{q\lbrack k\rbrack}} \right\}} & \left\lbrack {{Math}\mspace{14mu} {Figure}\mspace{14mu} 7} \right\rbrack \\\begin{matrix}{{i\lbrack k\rbrack} = {{1\mspace{14mu} {for}\mspace{14mu} k} = {0 \sim 19}}} \\{= {{{- 1}\mspace{14mu} {for}\mspace{14mu} k} = {20 \sim 59}}} \\{= {{1\mspace{14mu} {for}\mspace{14mu} k} = {60 \sim 115}}} \\{= {{{- 1}\mspace{14mu} {for}\mspace{14mu} k} = {116 \sim 187}}} \\{= {{1\mspace{14mu} {for}\mspace{14mu} k} = {188 \sim 291}}} \\{= {{{- 1}\mspace{14mu} {for}\mspace{14mu} k} = {292 \sim 427}}} \\{= {{1\mspace{14mu} {for}\mspace{14mu} k} = {428 \sim 627}}} \\{= {{{- 1}\mspace{14mu} {for}\mspace{14mu} k} = {628 \sim 891}}} \\{= {{1\mspace{14mu} {for}\mspace{14mu} k} = {892 \sim 1023}}}\end{matrix} & \; \\\begin{matrix}{{q\lbrack k\rbrack} = {{1\mspace{14mu} {for}\mspace{14mu} k} = {0 \sim 39}}} \\{= {{{- 1}\mspace{14mu} {for}\mspace{14mu} k} = {40 \sim 79}}} \\{= {{1\mspace{14mu} {for}\mspace{14mu} k} = {80 \sim 151}}} \\{= {{{- 1}\mspace{14mu} {for}\mspace{14mu} k} = {152 \sim 223}}} \\{= {{1\mspace{14mu} {for}\mspace{14mu} k} = {224 \sim 359}}} \\{= {{{- 1}\mspace{14mu} {for}\mspace{14mu} k} = {360 \sim 495}}} \\{= {{1\mspace{14mu} {for}\mspace{14mu} k} = {496 \sim 759}}} \\{= {{{- 1}\mspace{14mu} {for}\mspace{14mu} k} = {760 \sim 1023}}}\end{matrix} & \;\end{matrix}$

The binary chirp-like sequence can be represented by Math FIG. 7. Thesignal represented by Math FIG. 7 is an embodiment of the binarychirp-like sequence.

The binary chirp-like sequence is a sequence that is quantized such thatthe real-number part and imaginary part of each signal valueconstituting the above-described chirp-like sequence have only twovalues of ‘1’ and ‘−1’. The binary chirp-like sequence according toanother embodiment of the present invention can have the real-numberpart and imaginary part having only two signal values of ‘−0.707 (−1divided by square root of 2)’ and ‘0.707’(1 divided by square root of2). The quantized value of the real-number part and imaginary part ofthe binary chirp-like sequence can be changed by the designer. In MathFIG. 7, i[k] represents the real-number part of each signal constitutingthe sequence and q[k] represents the imaginary part of each signalconstituting the sequence.

The binary chirp-like sequence has the following advantages. Firstly,the binary chirp-like sequence does not generate dangerous delay sinceit is composed of signals having different periods. Secondly, the binarychirp-like sequence has correlation characteristic similar to guardinterval correlation and thus provides correct symbol timing informationcompared to conventional broadcast systems and has higher noiseresistance on a multi-path channel than a sequence having delta-likecorrelation characteristic such as m-sequence. Thirdly, when scramblingis performed using the binary chirp-like sequence, bandwidth is lessincreased compared to the original signal. Fourthly, since the binarychirp-like sequence is a binary level sequence, a receiver with reducedcomplexity can be designed when the binary chirp-like sequence is used.

In the graph showing the waveforms of the binary chirp-like sequence, asolid line represents a waveform corresponding to real-number parts anda dotted line represents a waveform corresponding to imaginary parts.Both the waveforms of the real-number parts and imaginary parts of thebinary chirp-like sequence correspond to a square wave, differently fromthe chirp-like sequence.

FIG. 39 is a graph showing a result obtained when a scrambling sequenceaccording to another embodiment of the present invention is used. Thegraph shows the waveform of signal c(t) output from the above-describedcorrelator block when the binary chirp-like sequence is used. In thegraph, the peak may be a correlation peak according to cyclic prefix.

As described above with reference to FIG. 31, the signaling sequenceinterleaving block 18100 included in the preamble insertion blockaccording to an embodiment of the present invention can interleave thesignaling sequences for transmitting the input signaling informationaccording to the signaling sequence selected by the signaling sequenceselection block 18000.

A description will be given of a method through which the signalingsequence interleaving block 18100 according to an embodiment of thepresent invention interleaves the signaling information in the frequencydomain of the preamble.

FIG. 40 illustrates a signaling information interleaving procedureaccording to an embodiment of the present invention.

The preamble according to an embodiment of the present invention,described above with reference to FIG. 17, can have a size of 1K symboland only 384 active carriers from among carriers constituting the 1Ksymbol can be used. The size of the preamble or the number of activecarriers used can be changed by the designer. The signalling datacarried in the preamble is composed of 2 signalling fields, namely S1and S2.

As shown in FIG. 40, the signaling information carried by the preambleaccording to an embodiment of the present invention can be transmittedthrough bit sequences of S1 and bit sequences of S2.

The bit sequences of S1 and the bit sequences of S2 according to anembodiment of the present invention represent signaling sequences thatcan be allocated to active carriers to respectively carry signalinginformation (or signaling fields) included in the preamble.

Specifically, S1 can carry 3-bit signaling information and can beconfigured in a structure in which a 64-bit sequence is repeated twice.In addition, S1 can be located before and after S2. S2 is a single256-bit sequence and can carry 4-bit signaling information. The bitsequences of S1 and S2 are represented as sequential numbers startingfrom 0 according to an embodiment of the present invention. Accordingly,the first bit sequence of S1 can be represented as S1(0) and the firstbit sequence of S2 can be represented as S2(0), as shown in FIG. 40.This can be changed by the designer.

S1 can carry information for identifying the signal frames included inthe super-frame described in FIG. 30, for example, a signal frameprocessed according to SISO, a signal frame processed according to MISOor information indicating FE. S2 can carry information about the FFTsize of the current signal frame, information indicating whether or notframes multiplexed in a super-frame are of the same type or the like.Information that can be carried by S1 and S2 can be changed according todesign.

As shown in FIG. 40, the signaling sequence interleaving block 18100according to an embodiment of the present invention can sequentiallyallocate S1 and S2 to active carriers corresponding to predeterminedpositions in the frequency domain.

In one embodiment of the present invention, 384 carriers are present andare represented′ as sequential numbers starting from 0. Accordingly, thefirst carrier according to an embodiment of the present invention can berepresented as a(0), as shown in FIG. 40. In FIG. 40, uncolored activecarriers are null carriers to which S1 or S2 is not allocated from amongthe 384 carriers.

As illustrated in FIG. 40, bit sequences of S1 can be allocated toactive carriers other than null carriers from among active carriers a(0)to a(63), bit sequences of S2 can be allocated to active carriers otherthan null carriers from among active carriers a(64) to a(319) and bitsequences of S1 can be allocated to active carriers other than nullcarriers from among active carriers a(320) to a(383).

According to the interleaving method illustrated in FIG. 40, theapparatus for receiving broadcast signals may not decode specificsignaling information affected by fading when frequency selective fadingoccurs due to multi-path interference and a fading period isconcentrated on a region to which the specific signaling information isallocated.

FIG. 41 illustrates a signaling information interleaving procedureaccording to another embodiment of the present invention.

According to the signaling information interleaving procedureillustrated in FIG. 41, the signaling information carried by thepreamble according to an embodiment of the present invention can betransmitted through bit sequences of S1, bit sequences of S2 and bitsequences of S3. The signalling data carried in the preamble is composedof 3 signalling fields, namely S1, S2 and S3.

As illustrated in FIG. 41, the bit sequences of S1, the bit sequences ofS2 and the bit sequences of S3 according to an embodiment of the presentinvention are signaling sequences that can be allocated to activecarriers to respectively carry signaling information (or signalingfields) included in the preamble.

Specifically, each of S1, S2 and S3 can carry 3-bit signalinginformation and can be configured in a structure in which a 64-bitsequence is repeated twice. Accordingly, 2-bit signaling information canbe further transmitted compared to the embodiment illustrated in FIG.40.

In addition, S1 and S2 can respectively carry the signaling informationdescribed in FIG. 40 and S3 can carry signaling information about aguard length (or guard interval length). Signaling information carriedby S1, S2 and S3 can be changed according to design.

As illustrated in FIG. 41, bit sequences of S1, S2 and S3 can berepresented as sequential numbers starting from 0, that is, S1(0), . . .. In the present embodiment of the invention, 384 carriers are presentand are represented as sequential numbers starting from 0, that is,b(0), . . . . This can be modified by the designer.

As illustrated in FIG. 41, S1; S2 and S3 can be sequentially andrepeatedly allocated to active carriers corresponding to predeterminedpositions in the frequency domain.

Specifically, bit sequences of S1, S2 and S3 can be sequentiallyallocated to active carriers other than null packets from among activecarriers b(0) to b(383) according to Math FIG. 8.

b(n)=S1(n/3) when n mod 3=0 and 0≦n<192

b(n)=S2((n−1)/3) when n mod 3=1 and 0≦n<192

b(n)=S3((n−2)/3) when n mod 3=2 and 0≦n<192

b(n)=S1((n−192)/3) when n mod 3=0 and 192≦n<384

b(n)=S2((n−192−1)/3) when n mod 3=1 and 192≦n<384

b(n)=S3((n−192−2)/3) when n mod 3=2 and 192≦n<384  [Math FIG. 8]

According to the interleaving method illustrated in FIG. 41, it ispossible to transmit a larger amount of signaling information than theinterleaving method illustrated in FIG. 40. Furthermore, even iffrequency selective fading occurs due to multi-path interference, theapparatus for receiving broadcast signals can uniformly decode signalinginformation since a fading period can be uniformly distributed in aregion to which signaling information is allocated.

FIG. 42 illustrates a signaling decoder according to an embodiment ofthe present invention.

The signaling decoder illustrated in FIG. 42 corresponds to anembodiment of the signaling decoder illustrated in FIG. 33 and caninclude a descrambler 27000, a demapper 27100, a signaling sequencedeinterleaver 27200 and a maximum likelihood detector 27300. Adescription will be given of operation of each block of the signalingdecoder.

The descrambler 27000 can descramble a signal output from the dataextractor. In this case, the descrambler 27000 can perform descramblingby multiplying the signal output from the data extractor by thescrambling sequence. The scrambling sequence according to an embodimentof the present invention can correspond to one of the sequencesdescribed with reference to FIGS. 21, 22, 23, 24 and 25.

The demapper 27100 can demap the signal output from the descrambler27000 to output sequences having a soft value.

The signaling sequence deinterleaver 27200 can rearrange uniformlyinterleaved sequences as consecutive sequences in the original order byperforming deinterleaving corresponding to a reverse process of theinterleaving process described in FIGS. 25 and 26.

The maximum likelihood detector 27300 can decode preamble signalinginformation using the sequences output from the signaling sequencedeinterleaver 27200.

FIG. 43 is a graph showing the performance of the signaling decoderaccording to an embodiment of the present invention.

The graph of FIG. 43 shows the performance of the signaling decoder asthe relationship between correct decoding probability and SNR in thecase of perfect synchronization, 1 sample delay, 0 dB and 270 degreesingle ghost.

Specifically, first, second and third curves 28000 respectively show thedecoding performance of the signaling decoder for S1, S2 and S3 when theinterleaving method illustrated in FIG. 40 is employed, that is, S1, S2and S3 are sequentially allocated to active carriers and transmitted.Fourth, fifth and sixth curves 28100 respectively show the decodingperformance of the signaling decoder for S1, S2 and S3 when theinterleaving method illustrated in FIG. 41 is employed, that is, S1, S2and S3 are sequentially allocated to active carriers corresponding topredetermined positions in the frequency domain in a repeated manner andtransmitted. Referring to FIG. 43, it can be known that there is a largedifference between signaling decoding performance for a regionconsiderably affected by fading and signaling decoding performance for aregion that is not affected by fading when a signal processed accordingto the interleaving method illustrated in FIG. 40 is decoded. When asignal processed according to the interleaving method illustrated inFIG. 41 is decoded, however, uniform signaling decoding performance isachieved for S1, S2 and S3.

FIG. 44 illustrates a preamble insertion block according to anotherembodiment of the present invention.

The preamble insertion block shown in FIG. 44 corresponds to anotherembodiment of the preamble insertion block 7500 illustrated in FIG. 11.

As shown in FIG. 44, the preamble insertion block can include a ReedMuller encoder 29000, a data formatter 29100, a cyclic delay block29200, an interleaver 29300, a DQPSK (differential quadrature phaseshift keying)/DBPSK (differential binary phase shift keying) mapper29400, a scrambler 29500, a carrier allocation block 29600, a carrierallocation table block 29700, an IFFT block 29800, a scrambled guardinsertion block 29900, a preamble repeater 29910 and a multiplexingblock 29920. Each block may be modified or may not be included in thepreamble insertion block according to design. A description will begiven of operation of each block of the preamble insertion block.

The Reed Muller encoder 29000 can receive signaling information to becarried by the preamble and perform Reed Muller encoding on thesignaling information. When Reed Muller encoding is performed,performance can be improved compared to signaling using an orthogonalsequence or signaling using the sequence described in FIG. 31.

The data formatter 29100 can receive bits of the signaling informationon which Reed Muller encoding has been performed and format the bits torepeat and arrange the bits.

The DQPSK/DBPSK mapper 29400 can map the formatted bits of the signalinginformation according to DQPSK or DBPSK and output the mapped signalinginformation.

When the DQPSK/DBPSK mapper 29400 maps the formatted bits of thesignaling information according to DBPSK, the operation of the cyclicdelay block 29200 can be omitted. The interleaver 29300 can receive theformatted bits of the signaling information and perform frequencyinterleaving on the formatted bits of the signaling information tooutput interleaved data. In this case, the operation of the interleavercan be omitted according to design.

When the DQPSK/DBPSK mapper 29400 maps the formatted bits of thesignaling information according to DQPSK, the data formatter 29100 canoutput the formatted bits of the signaling information to theinterleaver 29300 through path I shown in FIG. 44.

The cyclic delay block 29200 can perform cyclic delay on the formattedbits of the signaling information output from the data formatter 29100and then output the cyclic-delayed bits to the interleaver 29300 throughpath Q shown in FIG. 44. When cyclic Q-delay is performed, performanceon a frequency selective fading channel is improved.

The interleaver 29300 can perform frequency interleaving on thesignaling information received through paths I and Q and the cyclicQ-delayed signaling information to output interleaved information. Inthis case, the operation of the interleaver 29300 can be omittedaccording to design.

Math FIGS. 6 and 7 represent the relationship between input informationand output information or a mapping rule when the DQPSK/DBPSK mapper29400 maps the signaling information input thereto according to DQPSKand DBPSK.

As shown in FIG. 44, the input information of the DQPSK/DBPSK mapper29400 can be represented as si[in] and sq[n] and the output informationof the DQPSK/DBPSK mapper 29400 can be represented as mi[in] and mq[n].

m _(i)[−1]=1,

m _(i) [n]=m _(i) [n−1] if s _(i) [n]=0

m _(i) [n]=−m _(i) [n−1] if s _(i) [n]=1,

m _(q) [n]=0,n=0˜l, l: # of Reed Muller encoded signaling bits  [MathFIG. 9]

y[−1]=0

y[n]=y[n−1] if s _(i) [n]=0 and s _(q) [n]=0

y[n]=(y[n−1]+3)mod 4 if s _(i) [n]=0 and s _(q) [n]=1

y[n]=(y[n−1]+1)mod 4 if s _(i) [n]=1 and s _(q) [n]=0

y[n]=(y[n−1]+2)mod 4 if s _(i) [n]=1 and s _(q) [n]=1, n=0˜l, l: # ofReed Muller encoded signaling bits

m _(i) [n]=m _(q) [n]=if y[n]=0

m _(i) [n]=m _(q) [n]=if y[n]=1

m _(i) [n]=m _(q) [n]=if y[n]=2

m _(i) [n]=m _(q) [n]=if y[n]=3, n=0˜l, l# of Reed Muller encodedsignaling bits  [Math FIG. 10]

The scrambler 29500 can receive the mapped signaling information outputfrom the DQPSK/DBPSK mapper 29400 and multiply the signaling informationby the scrambling sequence.

The carrier allocation block 29600 can allocate the signalinginformation processed by the scrambler 29500 to predetermined carriersusing position information output from the carrier allocation tableblock 29700.

The IFFT block 29800 can transform the carriers output from the carrierallocation block 29600 into an OFDM signal in the time domain.

The scrambled guard insertion block 29900 can insert a guard intervalinto the OFDM signal to generate a preamble. The guard intervalaccording to one embodiment of the present invention can correspond tothe guard interval in the scrambled cyclic prefix form described in FIG.32 and can be generated according to the method described in FIG. 32.

The preamble repeater 29910 can repeatedly arrange the preamble in asignal frame. The preamble according to one embodiment of the presentinvention can have the preamble structure described in FIG. 32 and canbe transmitted through one signal frame only once.

When the preamble repeater 29910 repeatedly allocate the preamble withinone signal frame, the OFDM symbol region and scrambled cyclic prefixregion of the preamble can be separated from each other. The preamblecan include the scrambled cyclic prefix region and the OFDM symbolregion, as described above. In the specification, the preamblerepeatedly allocated by the preamble repeater 29910 can also be referredto as a preamble. The repeated preamble structure may be a structure inwhich the OFDM symbol region and the scrambled cyclic prefix region arealternately repeated. Otherwise, the repeated preamble structure may bea structure in which the OFDM symbol region is allocated, the scrambledprefix region is consecutively allocated twice or more and then the OFDMsymbol region is allocated. Furthermore, the repeated preamble structuremay be a structure in which the scrambled cyclic prefix region isallocated, the OFDM symbol region is consecutively allocated twice ormore and then the scrambled cyclic prefix region is allocated. Apreamble detection performance level can be controlled by adjusting thenumber of repetitions of the OFDM symbol region or scrambled cyclicprefix region and positions in which the OFDM symbol region andscrambled cyclic prefix region are allocated.

When the same preamble is repeated in one frame, the apparatus forreceiving broadcast signals can stably detect the preamble even in thecase of low SNR and decode the signaling information.

The multiplexing block 29920 can multiplex the signal output from thepreamble repeater 29910 and the signal c(t) output from the guardsequence insertion block 7400 illustrated in FIG. 7 to output an outputsignal p(t). The output signal p(t) can be input to the waveformprocessing block 7600 described in FIG. 7.

FIG. 45 illustrates a structure of signaling data in a preambleaccording to an embodiment of the present invention.

Specifically, FIG. 45 shows the structure of the signaling data carriedon the preamble according to an embodiment of the present invention inthe frequency domain.

As shown in FIG. 45, (a) and (b) illustrate an embodiment in which thedata formatter 29100 described in FIG. 44 repeats or allocates dataaccording to code block length of Reed Muller encoding performed by theReed Muller encoder 29000.

The data formatter 29100 can repeat the signaling information outputfrom the Reed Muller encoder 29000 such that the signaling informationcorresponds to the number of active carriers based on code block lengthor arrange the signaling information without repeating the same. (a) and(b) correspond to a case in which the number of active carriers is 384.

Accordingly, when the Reed Muller encoder 29000 performs Reed Mullerencoding of a 64-bit block, as shown in (a), the data formatter 29100can repeat the same data six times. In this case, if the first orderReed Muller code is used in Reed Muller encoding, the signaling data maybe 7 bits.

When the Reed Muller encoder 29000 performs Reed Muller encoding of a256-bit block, as shown in (b), the data formatter 29100 can repeatformer 128 bits or later 124 bits of the 256-bit code block or repeat128 even-numbered bits or 124 odd-numbered bits. In this case, if thefirst order Reed Muller code is used in Reed Muller encoding, thesignaling data may be 8 bits.

As described above with reference to FIG. 44, the signaling informationformatted by the data formatter 29100 can be processed by the cyclicdelay block 29200 and the interleaver 29300 or mapped by the DQPSK/DBPSKmapper 29400 without being processed by the cyclic delay block 29200 andthe interleaver 29300, scrambled by the scrambler 29500 and input to thecarrier allocation block 29600.

As shown in FIG. 45, (c) illustrates a method of allocating thesignaling information to active carriers in the carrier allocation block29600 according to one embodiment. As shown in (c), b(n) representscarriers to which data is allocated and the number of carriers can be384 in one embodiment of the present invention. Colored carriers fromamong the carriers shown in (c) refer to active carriers and uncoloredcarriers refer to null carriers. The positions of the active carriersillustrated in FIG. 45-(c) can be changed according to design.

FIG. 46 illustrates a procedure of processing signaling data carried ona preamble according to one embodiment.

The signaling data carried on a preamble may include a plurality ofsignaling sequences. Each signaling sequence may be 7 bits. The numberand size of signaling sequences can be changed by the designer.

In the figure, (a) illustrates a signaling data processing procedureaccording to an embodiment when the signaling data carried on thepreamble is 14 bits. In this case, the signaling data carried on thepreamble can include two signaling sequences which are respectivelyreferred to as signaling 1 and signaling 2. Signaling 1 and signaling 2may correspond to the above-described signaling sequences S1 and S2.

Each of signaling 1 and signaling 2 can be encoded into a 64-bit ReedMuller code by the above-described Reed Muller encoder. In the figure,(a) illustrates Reed Muller encoded signaling sequence blocks 32010 and32040.

The signaling sequence blocks 32010 and 32040 of the encoded signaling 1and signaling 2 can be repeated three times by the above-described dataformatter. In the figure, (a) illustrates repeated signaling sequenceblocks 32010, 32020 and 32030 of signaling 1 and repeated signalingsequence blocks 32040, 32050 and 32060 of repeated signaling 2. Since aReed-Muller encoded signaling sequence block is 64 bits, each of thesignaling sequence blocks of signaling 1 and signaling 2, which arerepeated three times, is 192 bits.

Signaling 1 and signaling 2 composed of 6 blocks 32010, 32020, 32030,32040, 32050 and 32060 can be allocated to 384 carriers by theabove-described carrier allocation block. In the figure (a), b(0) is thefirst carrier and b(1) and b(2) are carriers. 384 carriers b(0) tob(383) are present in one embodiment of the present invention. Coloredcarriers from among the carriers shown in the figure refer to activecarriers and uncolored carriers refer to null carriers. The activecarrier represents a carrier to which signaling data is allocated andthe null carrier represents a carrier to which signaling data is notallocated. In this specification, active carrier can also be referred toas a carrier. Data of signaling 1 and data of signaling 2 can bealternately allocated to carriers. For example, the data of signaling 1can be allocated to b(0), the data of signaling 2 can be allocated tob(7) and the data of signaling 1 can be allocated to b(24). Thepositions of the active carriers and null carriers can be changed by thedesigner.

In the figure, (b) illustrates a signaling data processing procedurewhen the signaling data transmitted through the preamble is 21 bits. Inthis case, the signaling data transmitted through the preamble caninclude three signaling sequences which are respectively referred to assignaling 1, signaling 2 and signaling 3. Signaling 1, signaling 2 andsignaling 3 may correspond to the above-described signaling sequencesS1, S2 and S3.

Each of signaling 1, signaling 2 and signaling 3 can be encoded into a64-bit Reed-Muller code by the above-described Reed-Muller encoder. Inthe figure, (b) illustrates Reed-Muller encoded signaling sequenceblocks 32070, 32090 and 32110.

The signaling sequence blocks 32070, 32090 and 32110 of the encodedsignaling 1, signaling 2 and signaling 3 can be repeated twice by theabove-described data formatter. In the figure, (b) illustrates therepeated signaling sequence blocks 32070 and 32080 of signaling 1,repeated signaling sequence blocks 32090 and 32100 of signaling 2 andrepeated signaling sequence blocks 32110 and 32120 of signaling 3. Sincea Reed-Muller encoded signaling sequence block is 64 bits, each of thesignaling sequence blocks of signaling 1, signaling 2 and signaling 3,which are repeated twice, is 128 bits.

Signaling 1, signaling 2 and signaling 3 composed of 6 blocks 32070,32080, 32090, 32100, 32110 and 32120 can be allocated to 384 carriers bythe above-described carrier allocation block. In the figure (b), b(0) isthe first carrier and b(1) and b(2) are carriers. 384 carriers b(0) tob(383) are present in one embodiment of the present invention. Coloredcarriers from among the carriers shown in the figure refer to activecarriers and uncolored carriers refer to null carriers. The activecarrier represents a carrier to which signaling data is allocated andthe null carrier represents a carrier to which signaling data is notallocated. Data of signaling 1, signaling 2 and data of signaling 3 canbe alternately allocated to carriers. For example, the data of signaling1 can be allocated to b(0), the data of signaling 2 can be allocated tob(7), the data of signaling 3 can be allocated to b(24) and the data ofsignaling 1 can be allocated to b(31). The positions of the activecarriers and null carriers can be changed by the designer.

As illustrated in (a) and (b) of the figure, trade off between signalingdata capacity and signaling data protection level can be achieved bycontrolling the length of an FEC encoded signaling data block. That is,when the signaling data block length increases, signaling data capacityincreases whereas the number of repetitions by the data formatter andthe signaling data protection level decrease. Accordingly, varioussignaling capacities can be selected.

FIG. 47 illustrates a preamble structure repeated in the time domainaccording to one embodiment.

As described above, the preamble repeater can alternately repeat dataand a scrambled guard interval. In the following description, a basicpreamble refers to a structure in which a data region follows ascrambled guard interval.

In the figure, (a) illustrates a structure in which the basic preambleis repeated twice in a case in which the preamble length is 4N. Since apreamble having the structure of (a) includes the basic preamble, thepreamble can be detected even by a normal receiver in an environmenthaving a high signal-to-noise ratio (SNR) and detected using therepeated structure in an environment having a low SNR. The structure of(a) can improve decoding performance of the receiver since signalingdata is repeated in the structure.

In the figure, (b) illustrates a preamble structure when the preamblelength is 5N. The structure of (b) is started with data and then a guardinterval and data are alternately allocated. This structure can improvepreamble detection performance and decoding performance of the receiversince the data is repeated a larger number of times (3N) than thestructure of (a).

In the figure, (c) illustrates a preamble structure when the preamblelength is 5N. Distinguished from the structure of (b), the structure of(c) is started with the guard interval and then the data and the guardinterval are alternately allocated. The structure of (c) has a smallernumber (2N) of repetitions of data than the structure of (b) althoughthe preamble length is identical to that of the structure of (b), andthus the structure of (c) may deteriorate decoding performance of thereceiver. However, the preamble structure of (c) has an advantage that aframe is started in the same manner as a normal frame since the dataregion follows the scrambled guard interval.

FIG. 48 illustrates a preamble detector and a correlation detectorincluded in the preamble detector according to an embodiment of thepresent invention.

FIG. 48 illustrates an embodiment of the above-described preambledetector for the preamble structure of (b) in the above-described figureshowing the preamble structure repeated in the time domain.

The preamble detector according to the present embodiment can include acorrelation detector 34010, an FFT block 34020, an ICFO estimator 34030,a data extractor 34040 and/or a signaling decoder 34050.

The correlation detector 34010 can detect a preamble. The correlationdetector 34010 can include two branches. The above-described repeatedpreamble structure can be a structure in which the scrambled guardinterval and data region are alternatively assigned. Branch 1 can beused to obtain correlation of a period in which the scrambled guardinterval is located prior to the data region in the preamble. Branch 2can be used to obtain correlation of a period in which the data regionis located prior to the scrambled guard interval in the preamble.

In the preamble structure of (b) in the above figure showing thepreamble structure repeated in the time domain, in which the data regionand scrambled guard interval are repeated, the period in which thescrambled guard interval is located prior to the data region appearstwice and the period in which the data region is located prior to thescrambled guard interval appears twice. Accordingly, 2 correlation peakscan be generated in each of branch 1 and branch 2. The 2 correlationbranches generated in each branch can be summed. A correlator includedin each branch can correlate the summed correlation peak with ascrambling sequence. The correlated peaks of branch 1 and branch 2 canbe summed and a peak detector can detect the preamble position from thesummed peak of branch 1 and branch 2 and perform OFDM symbol timingsynchronization and fractional frequency offset synchronization.

The FFT block 34020, ICFO estimator 34030, data extractor 34040 andsignaling decoder 34050 can operate in the same manner as theabove-described corresponding blocks.

FIG. 49 illustrates a preamble detector according to another embodimentof the present invention.

The preamble detector shown in FIG. 49 corresponds to another embodimentof the preamble detector 9300 described in FIGS. 9 and 20 and canperform operation corresponding to the preamble insertion blockillustrated in FIG. 44.

As shown in FIG. 49, the preamble detector according to anotherembodiment of the present invention can include a correlation detector,an FFT block, an ICFO estimator, a carrier allocation table block, adata extractor and a signaling decoder 31100 in the same manner as thepreamble detector described in FIG. 33. However, the preamble detectorshown in FIG. 49 is distinguished from the preamble detector shown inFIG. 33 in that the preamble detector shown in FIG. 49 includes apreamble combiner 31000. Each block may be modified or omitted from thepreamble detector according to design.

Description of the same blocks as those of the preamble detectorillustrated in FIG. 33 is omitted and operations of the preamblecombiner 31000 and signaling decoder 31100 are described.

The preamble combiner 31000 can include n delay blocks 31010 and anadder 31020. The preamble combiner 31000 can combine received signals toimprove signal characteristics when the preamble repeater 29910described in FIG. 44 repeatedly allocate the same preamble to one signalframe.

As shown in FIG. 49, the n delay blocks 31010 can delay each preamble byp*n−1 in order to combine repeated preambles. In this case, p representsa preamble length and n represents the number of repetitions.

The adder 31020 can combine the delayed preambles.

The signaling decoder 31100 corresponds to another embodiment of thesignaling decoder illustrated in FIG. 42 and can perform reverseoperations of the operations of the Reed Muller encoder 29000, dataformatter 29100, cyclic delay block 29200, interleaver 29300,DQPSK/DBPSK mapper 29400 and scrambler 29500 included in the preambleinsertion block illustrated in FIG. 44.

As shown in FIG. 49, the signaling decoder 31100 can include adescrambler 31110, a differential decoder 31120, a deinterleaver 31130,a cyclic delay block 31140, an I/Q combiner 31150, a data deformatter31160 and a Reed Muller decoder 31170.

The descrambler 31110 can descramble a signal output from the dataextractor.

The differential decoder 31120 can receive the descrambled signal andperform DBPSK or DQPSK demapping on the descrambled signal.

Specifically, when a signal on which DQPSK mapping has been performed inthe apparatus for transmitting broadcast signals is received, thedifferential decoder 31120 can phase-rotate a differential-decodedsignal by π/4. Accordingly, the differential decoded signal can bedivided into in-phase and quadrature components.

If the apparatus for transmitting broadcast signals has performedinterleaving, the deinterleaver 31130 can deinterleave the signal outputfrom the differential decoder 31120.

If the apparatus for transmitting broadcast signals has performed cyclicdelay, the cyclic delay block 31140 can perform a reverse process ofcyclic delay.

The I/Q combiner 31150 can combine I and Q components of thedeinterleaved or delayed signal.

If a signal on which DBPSK mapping has been performed in the apparatusfor transmitting broadcast signals is received, the I/Q combiner 31150can output only the I component of the deinterleaved signal.

The data deformatter 31160 can combine bits of signals output from theI/Q combiner 31150 to output signaling information. The Reed Mullerdecoder 31170 can decode the signaling information output from the datadeformatter 31160.

Accordingly, the apparatus for receiving broadcast signals according toan embodiment of the present invention can acquire the signalinginformation carried by the preamble through the above-describedprocedure.

FIG. 50 illustrates a preamble detector and a signaling decoder includedin the preamble detector according to an embodiment of the presentinvention.

FIG. 50 shows an embodiment of the above-described preamble detector.

The preamble detector according to the present embodiment can include acorrelation detector 36010, an FFT block 36020, an ICFO estimator 36030,a data extractor 36040 and/or a signaling decoder 36050.

The correlation detector 36010, FFT block 36020, ICFO estimator 36030and data extractor 36040 can perform the same operations as those of theabove-described corresponding blocks.

The signaling decoder 36050 can decode the preamble. The signalingdecoder 36050 according to the present embodiment can include a dataaverage module 36051, a descrambler 36052, a differential decoder 36053,a deinterleaver 36054, a cyclic delay 36055, an I/Q combiner 36056, adata deformatter 36057 and/or a Reed-Muller decoder 36058.

The data average module 36051 can calculate the average of repeated datablocks to improve signal characteristics when the preamble has repeateddata blocks. For example, if a data block is repeated three times, asillustrated in (b) of the above figure showing the preamble structurerepeated in the time domain, the data average module 36051 can calculatethe average of the 3 data blocks to improve signal characteristics. Thedata average module 36051 can output the averaged data to the nextmodule.

The descrambler 36052, differential decoder 36053, deinterleaver 36054,cyclic delay 36055, I/Q combiner 36056, data deformatter 36057 and ReedMuller decoder 36058 can perform the same operations as those of theabove-described corresponding blocks.

FIG. 51 is a view illustrating a frame structure of a broadcast systemaccording to an embodiment of the present invention.

The above-described cell mapper included in the frame structure modulemay locate cells for transmitting input SISO, MISO or MIMO processed DPdata, cells for transmitting common DP data, and cells for transmittingPLS data in a signal frame according to scheduling information. Then,the generated signal frames may be sequentially transmitted.

A broadcast signal transmission apparatus and transmission methodaccording to an embodiment of the present invention may multiplex andtransmit signals of different broadcast transception systems within thesame RF channel, and a broadcast signal reception apparatus andreception method according to an embodiment of the present invention maycorrespondingly process the signals. Thus, a broadcast signaltransception system according to an embodiment of the present inventionmay provide a flexible broadcast transception system.

Therefore, the broadcast signal transmission apparatus according to anembodiment of the present invention may sequentially transmit aplurality of superframes delivering data related to broadcast service.

FIG. 51(a) illustrates a superframe according to an embodiment of thepresent invention, and FIG. 51(b) illustrates the configuration of thesuperframe according to an embodiment of the present invention. Asillustrated in FIG. 51(b), the superframe may include a plurality ofsignal frames and a non-compatible frame (NCF). According to anembodiment of the present invention, the signal frames are time divisionmultiplexing (TDM) signal frames of a physical layer end, which aregenerated by the above-described frame structure module, and the NCF isa frame which is usable for a new broadcast service system in thefuture.

The broadcast signal transmission apparatus according to an embodimentof the present invention may multiplex and transmit various services,e.g., UHD, Mobile and MISO/MIMO, on a frame basis to simultaneouslyprovide the services in an RE. Different broadcast services may requiredifferent reception environments, transmission processes, etc. accordingto characteristics and purposes of the broadcast services.

Accordingly, different services may be transmitted on a signal framebasis, and the signal frames can be defined as different frame typesaccording to services transmitted therein. Further, data included in thesignal frames can be processed using different transmission parameters,and the signal frames can have different FFT sizes and guard intervalsaccording to broadcast services transmitted therein.

Accordingly, as illustrated in FIG. 51(b), the different-type signalframes for transmitting different services may be multiplexed using TDMand transmitted within a superframe.

According to an embodiment of the present invention, a frame type may bedefined as a combination of an FFT mode, a guard interval mode and apilot pattern, and information about the frame type may be transmittedusing a preamble portion within a signal frame. A detailed descriptionthereof will be given below.

Further, configuration information of the signal frames included in thesuperframe may be signaled through the above-described PLS, and may varyon a superframe basis.

FIG. 51(c) is a view illustrating the configuration of each signalframe. The signal frame may include a preamble, head/tail edge symbolsE_(H)/E_(T), one or more PLS symbols and a plurality of data symbols.This configuration is variable according to the intention of a designer.

The preamble is located at the very front of the signal frame and maytransmit a basic transmission parameter for identifying a broadcastsystem and the type of signal frame, information for synchronization,etc. Thus, the broadcast signal reception apparatus according to anembodiment of the present invention may initially detect the preamble ofthe signal frame, identify the broadcast system and the frame type, andselectively receive and decode a broadcast signal corresponding to areceiver type.

The head/tail edge symbols may be located after the preamble of thesignal frame or at the end of the signal frame. In the presentinvention, an edge symbol located after the preamble may be called ahead edge symbol and an edge symbol located at the end of the signalframe may be called a tail edge symbol. The names, locations or numbersof the edge symbols are variable according to the intention of adesigner. The head/tail edge symbols may be inserted into the signalframe to support the degree of freedom in design of the preamble andmultiplexing of signal frames having different frame types. The edgesymbols may include a larger number of pilots compared to the datasymbols to enable frequency-only interpolation and time interpolationbetween the data symbols. Accordingly, a pilot pattern of the edgesymbols has a higher density than that of the pilot pattern of the datasymbols.

The PLS symbols are used to transmit the above-described. PLS data andmay include additional system information (e.g., networktopology/configuration, PAPR use, etc.), frame type ID/configurationinformation, and information necessary to extract and decode DPs.

The data symbols are used to transmit DP data, and the above-describedcell mapper may locate a plurality of DPs in the data symbols.

A description is now given of DPs according to an embodiment of thepresent invention.

FIG. 52 is a view illustrating DPs according to an embodiment of thepresent invention.

As described above, data symbols of a signal frame may include aplurality of DPs. According to an embodiment of the present invention,the DPs may be divided into type 1 to type 3 according to mapping modes(or locating modes) in the signal frame.

FIG. 52(a) illustrates type1 DPs mapped to the data symbols of thesignal frame, FIG. 52(b) illustrates type2 DPs mapped to the datasymbols of the signal frame, and FIG. 52(c) illustrates type3 DPs mappedto the data symbols of the signal frame. FIGS. 52(a) to 52(c) illustrateonly a data symbol portion of the signal frame, and a horizontal axisrefers to a time axis while a vertical axis refers to a frequency axis.A description is now given of the type1 to type3 DPs.

As illustrated in FIG. 52(a), the type1 DPs refer to DPs mapped usingTDM in the signal frame.

That is, when the type1 DPs are mapped to the signal frame, a framestructure module (or cell mapper) according to an embodiment of thepresent invention may map corresponding DP cells in a frequency axisdirection. Specifically, the frame structure module (or cell mapper)according to an embodiment of the present invention may map cells of DP0in a frequency axis direction and, if an OFDM symbol is completelyfilled, move to a next OFDM symbol to continuously map the cells of DP0in a frequency axis direction. After the cells of DP0 are completelymapped, cells of DP1 and DP2 may also be mapped to the signal frame inthe same manner. In this case, the frame structure module (or cellmapper) according to an embodiment of the present invention may map thecells with an arbitrary interval between DPs.

Since the cells of the type1 DPs are mapped with the highest density onthe time axis, compared to other-type DPs, the type1 DPs may minimize anoperation time of a receiver. Accordingly, the type1 DPs are appropriateto provide a corresponding service to a broadcast signal receptionapparatus which should preferentially consider power saving, e.g., ahandheld or portable device which operates using a battery.

As illustrated in FIG. 52(b), the type2 DPs refer to DPs mapped usingfrequency division multiplexing (FDM) in the signal frame.

That is, when the type2 DPs are mapped to the signal frame, the framestructure module (or cell mapper) according to an embodiment of thepresent invention may map corresponding DP cells in a time axisdirection. Specifically, the frame structure module (or cell mapper)according to an embodiment of the present invention may preferentiallymap cells of DP0 on the time axis at a first frequency of an OFDMsymbol. Then, if the cells of DP0 are mapped to the last OFDM symbol ofthe signal frame on the time axis, the frame structure module (or cellmapper) according to an embodiment of the present invention maycontinuously map the cells of DP0 in the same manner from a secondfrequency of a first OFDM symbol.

Since the cells of the type2 DPs are transmitted with the widestdistribution in time, compared to other-type DPs, the type2 DPs areappropriate to achieve time diversity. However, since an operation timeof a receiver to extract the type2 DPs is longer than that to extractthe type1 DPs, the type2 DPs may not easily achieve power saving.Accordingly, the type2 DPs are appropriate to provide a correspondingservice to a fixed broadcast signal reception apparatus which stablyreceives power supply.

Since cells of each type2 DP are concentrated on a specific frequency, areceiver in a frequency selective channel environment may have problemto receive a specific DP.

Accordingly, after cell mapping, if frequency interleaving is applied ona symbol basis, frequency diversity may be additionally achieved andthus the above-described problem may be solved.

As illustrated in FIG. 52(c), the type3 DPs correspond to anintermediate form between the type1 DPs and the type2 DPs and refer toDPs mapped using time & frequency division multiplexing (TFDM) in thesignal frame.

When the type3 DPs are mapped to the signal frame, the frame structuremodule (or cell mapper) according to an embodiment of the presentinvention may equally partition the signal frame, define each partitionas a slot, and map cells of corresponding DPs in a time axis directionalong the time axis only within the slot.

Specifically, the frame structure module (or cell mapper) according toan embodiment of the present invention may preferentially map cells ofDP0 on the time axis at a first frequency of a first OFDM symbol. Then,if the cells of DP0 are mapped to the last OFDM symbol of the slot onthe time axis, the frame structure module (or cell mapper) according toan embodiment of the present invention may continuously map the cells ofDP0 in the same manner from a second frequency of the first OFDM symbol.

In this case, a trade-off between time diversity and power saving ispossible according to the number and length of slots partitioned fromthe signal frame. For example, if the signal frame is partitioned into asmall number of slots, the slots have a large length and thus timediversity may be achieved as in the type2 DPs. If the signal frame ispartitioned into a large number of slots, the slots have a small lengthand thus power saving may be achieved as in the type1 DPs.

FIG. 53 is a view illustrating type1 DPs according to an embodiment ofthe present invention.

FIG. 53 illustrates an embodiment in which the type1 DPs are mapped to asignal frame according to the number of slots. Specifically, FIG. 53(a)shows a result of mapping the type1 DPs when the number of slots is 1,and FIG. 53(b) shows a result of mapping the type1 DPs when the numberof slots is 4.

To extract cells of each DP mapped in the signal frame, the broadcastsignal reception apparatus according to an embodiment of the presentinvention needs type information of each DP and signaling information,e.g., DP start address information indicating an address to which afirst cell of each DP is mapped, and FEC block number information ofeach DP allocated to a signal frame.

Accordingly, as illustrated in FIG. 53(a), the broadcast signaltransmission apparatus according to an embodiment of the presentinvention may transmit signaling information including DP start addressinformation indicating an address to which a first cell of each DP ismapped (e.g., DP0_St, DP1_St, DP2_St, DP3_St, DP4_St), etc.

FIG. 53(b) shows a result of mapping the type1 DPs when the signal frameis partitioned into 4 slots. Cells of DPs mapped to each slot may bemapped in a frequency direction. As described above, if the number ofslots is large, since cells corresponding to a DP are mapped anddistributed with a certain interval, time diversity may be achieved.However, since the number of cells of a DP mapped to a single signalframe is not always divided by the number of slots, the number of cellsof a DP mapped to each slot may vary. Accordingly, if a mapping rule isestablished in consideration of this, an address to which a first cellof each DP is mapped may be an arbitrary location in the signal frame. Adetailed description of the mapping method will be given below. Further,when the signal frame is partitioned into a plurality of slots, thebroadcast signal reception apparatus needs information indicating thenumber of slots to obtain cells of a corresponding DP. In the presentinvention, the information indicating the number of slots may beexpressed as N_Slot. Accordingly, the number of slots of the signalframe of FIG. 53(a) may be expressed as N_Slot=1 and the number of slotsof the signal frame of FIG. 53(b) may be expressed as N_Slot=4.

FIG. 54 is a view illustrating type2 DPs according to an embodiment ofthe present invention.

As described above, cells of a type2 DP are mapped in a time axisdirection and, if the cells of the DP are mapped to the last OFDM symbolof a signal frame on a time axis, the cells of the DP may becontinuously mapped in the same manner from a second frequency of afirst OFDM symbol.

As described above in relation to FIG. 53, even in the case of the type2DPs, to extract cells of each DP mapped in the signal frame, thebroadcast signal reception apparatus according to an embodiment of thepresent invention needs type information of each DP and signalinginformation, e.g., DP start address information indicating an address towhich a first cell of each DP is mapped, and FEC block numberinformation of each DP allocated to a signal frame.

Accordingly, as illustrated in FIG. 54, the broadcast signaltransmission apparatus according to an embodiment of the presentinvention may transmit DP start address information indicating anaddress to which a first cell of each DP is mapped (e.g., DP0_St,DP1_St, DP2_St, DP3_St, DP4_St). Further, FIG. 54 illustrates a case inwhich the number of slots is 1, and the number of slots of the signalframe of FIG. 54 may be expressed as N_Slot=1.

FIG. 55 is a view illustrating type3 DPs according to an embodiment ofthe present invention.

The type3 DPs refer to DPs mapped using TFDM in a signal frame asdescribed above, and may be used when power saving is required whilerestricting or providing time diversity to a desired level. Like thetype2 DPs, the type3 DPs may achieve frequency diversity by applyingfrequency interleaving on an OFDM symbol basis.

FIG. 55(a) illustrates a signal frame in a case when a DP is mapped to aslot, and FIG. 55(b) illustrates a signal frame in a case when a DP ismapped to two or more slots. Both FIGS. 55(a) and 55(b) illustrate acase in which the number of slots is 4, and the number of slots of thesignal frame may be expressed as N_Slot=4.

Further, as illustrated in FIGS. 18 and 19, the broadcast signaltransmission apparatus according to an embodiment of the presentinvention may transmit DP start address information indicating anaddress to which a first cell of each DP is mapped (e.g., DP0_St,DP1_St, DP2_St, DP3_St, DP4_St).

In FIG. 55(b), time diversity different from that achieved in FIG. 55(a)may be achieved. In this case, additional signaling information may beneeded.

As described above in relation to FIGS. 18 to 20, the broadcast signaltransmission apparatus according to an embodiment of the presentinvention may transmit signaling information including DP start addressinformation indicating an address to which a first cell of each DP ismapped (e.g., DP0_St, DP1_St, DP2_St, DP3_St, DP4_St), etc. In thiscase, the broadcast signal transmission apparatus according to anembodiment of the present invention may transmit only the start addressinformation of DP0 which is initially mapped, and transmit an offsetvalue based on the start address information of DP0 for the other DPs.If the DPs are equally mapped, since mapping intervals of the DPs arethe same, a receiver may achieve start locations of the DPs usinginformation about a start location of an initial DP, and an offsetvalue. Specifically, when the broadcast signal transmission apparatusaccording to an embodiment of the present invention transmits offsetinformation having a certain size based on the start address informationof DP0, the broadcast signal reception apparatus according to anembodiment of the present invention may calculate a start location ofDP1 by adding the above-described offset information to the startaddress information of DP0. In the same manner, the broadcast signalreception apparatus according to an embodiment of the present inventionmay calculate a start location of DP2 by adding the above-describedoffset information twice to the start address information of DP0. If theDPs are not equally mapped, the broadcast signal transmission apparatusaccording to an embodiment of the present invention may transmit thestart address information of DP0 and offset values (OFFSET 1, OFFSET 2,. . . ) indicating intervals of the other DPs based on the startlocation of DP0. In this case, the offset values may be the same ordifferent. Further, the offset value(s) may be included and transmittedin PLS signaling information or in-band signaling information to bedescribed below with reference to FIG. 68. This is variable according tothe intention of a designer.

A description is now given of a method for mapping a DP using resourceblocks (RBs) according to an embodiment of the present invention.

An RB is a certain unit block for mapping a DP and may be called a datamapping unit in the present invention. RB based resource allocation isadvantageous in intuitively and easily processing DP scheduling andpower saving control. According to an embodiment of the presentinvention, the name of the RB is variable according to the intention ofa designer and the size of RB may be freely set within a range whichdoes not cause a problem in bit-rate granularity.

The present invention may exemplarily describe a case in which the sizeof RB is a value obtained by multiplying or dividing the number ofactive carriers (NoA) capable of transmitting actual data in an OFDMsymbol, by an integer. This is variable according to the intention of adesigner. If the RB has a large size, resource allocation may besimplified. However, the size of RB indicates a minimum unit of anactually supportable bit rate and thus should be determined withappropriate consideration.

FIG. 56 is a view illustrating RBs according to an embodiment of thepresent invention.

FIG. 56 illustrates an embodiment in which DP0 is mapped to a signalframe using RBs when the number of FEC blocks of DP0 is 10. A case inwhich the length of LDPC blocks is 64K and a QAM modulation value is256QAM as transmission parameters of DP0, a FFT mode of the signal frameis 32K, and a scattered pilot pattern is PP32-2 (i.e., the interval ofpilots delivering carriers is Dx=32, and the number of symbols includedin a scattered pilot sequence is Dy=2) is described as an example. Inthis case, the size of FEC block corresponds to 8100 cells, and NoA canbe assumed as 27584. Assuming that the size of RB is a value obtained bydividing NoA by 4, the size of RB corresponds to 6896 cells and may beexpressed as L_RB=NoA/4.

In this case, when the size of FEC blocks and the size of RBs arecompared on a cell basis, a relationship of the size of 10×FECblocks=the size of 11×RBs+5144 cells is established. Accordingly, to mapthe 10 FEC blocks to a single signal frame on an RB basis, the framestructure module (or cell mapper) according to an embodiment of thepresent invention may map data of the 10 FEC blocks sequentially to the11 RBs to map the 11 RBs to a current signal frame, and map theremaining data corresponding to the 5144 cells to a next signal frametogether with next FEC blocks.

FIG. 57 is a view illustrating a procedure for mapping RBs to framesaccording to an embodiment of the present invention.

Specifically, FIG. 57 illustrates a case in which contiguous signalframes are transmitted.

When a variable bit rate is supported, each signal frame may have adifferent number of FEC blocks transmittable therein.

FIG. 57(a) illustrates a case in which the number of FEC blocks to betransmitted in signal frame N is 10, a case in which the number of FECblocks to be transmitted in signal frame N+1 is 9, and a case in whichthe number of FEC blocks to be transmitted in signal frame N+2 is 11.

FIG. 57(b) illustrates a case in which the number of RB to be mapped tosignal frame N is 11, a case in which the number of RB to be mapped tosignal frame N+1 is 11, and a case in which the number of RB to bemapped to signal frame N+2 is 13.

FIG. 57(c) shows a result of mapping the RBs to signal frame N, signalframe N+1 and signal frame N+2.

As illustrated in FIGS. 22(a) and 22(b), when the number of FEC blocksto be transmitted in signal frame N is 10, since the size of 10 FECblocks equals to a value obtained by adding 5144 cells to the size of 11RBs, the 11 RBs may be mapped to and transmitted in signal frame N asillustrated in FIG. 57(c).

In addition, as illustrated in the center of FIG. 57(b), the remaining5144 cells form an initial part of a first RB among 11 RBs to be mappedto signal frame N+1. Accordingly, since a relationship of 5144 cells+thesize of 9 FEC blocks=the size of 11 RBs+2188 cells is established, 11RBs are mapped to and transmitted in signal frame N+1 and the remaining2188 cells form an initial part of a first RB among 13 RBs to be mappedto signal frame N+2. In the same manner, since a relationship of 2188cells+the size of 11 FEC blocks=the size of 13 RBs+1640 cells isestablished, 13 RBs are mapped to and transmitted in signal frame N+2and the remaining 1640 cells are mapped to and transmitted in a nextsignal frame. The size of FEC blocks is not the same as the size of NoAand thus dummy cells can be inserted. However, according to the methodillustrated in FIG. 57, there is no need to insert dummy cells and thusactual data may be more efficiently transmitted: Further, timeinterleaving or processing similar thereto may be performed on RBs to bemapped to a signal frame before the RBs are mapped to the signal frameand This is variable according to the intention of a designer.

A description is now given of a method of mapping DPs to a signal frameon an RB basis according to the above-described types of the DPs.

Specifically, in the present invention, the RB mapping method isdescribed by separating a case in which a plurality of DPs are allocatedto all available RBs in a signal frame from a case in which the DPs areallocated to only some RBs. The present invention may exemplarilydescribe a case in which the number of DPs is 3, the number of RBs in asignal frame is 80, and the size of RB is a value obtained by dividingNoA by 4. This case may be expressed as follows.

Number of DPs, N_DP=3

Number of RBs in a signal frame, N_RB=80

Size of RB, L_RB=NoA/4

Further, the present invention may exemplarily describe a case in whichDP0 fills 31 RBs, DP1 fills 15 RBs, and DP2 fills 34 RBs, as the case inwhich a plurality of DPs (DP0, DP1, DP2) are allocated to available RBsin a signal frame. This case may be expressed as follows.

{DP0, DP1, DP2}={31, 15, 34}

In addition, the present invention may exemplarily describe a case inwhich DP0 fills 7 RBs, DP1 fills 5 RBs, and DP2 fills 6 RBs, as the casein which a plurality of DPs (DP0, DP1, DP2) are allocated to only someRBs in a signal frame. This case may be expressed as follows.

{DP0, DP1, DP2}={7, 5, 6}

FIGS. 23 to 25 illustrate RB mapping according to the types of DPs.

The present invention may exemplarily define the following values todescribe an RB mapping rule according to the type of each DP.

L_Frame: Number of OFDM symbols in a signal frame

N_Slot: Number of slots in a signal frame

L_Slot: Number of OFDM symbols in a slot

N_RB_Sym: Number of RBs in an OFDM symbol

N_RB: Number of RBs in a signal frame

FIG. 58 is a view illustrating RB mapping of type1 DPs according to anembodiment of the present invention.

FIG. 58 illustrates a single signal frame, and a horizontal axis refersto a time axis while a vertical axis refers to a frequency axis. Acolored block located at the very front of the signal frame on the timeaxis corresponds to a preamble and signaling portion. As describedabove, according to an embodiment of the present invention, a pluralityof DPs may be mapped to a data symbol portion of the signal frame on aRB basis.

The signal frame illustrated in FIG. 58 consists of 20 OFMD symbols(L_Frame=20) and includes 4 slots (N_Slot=4). Further, each slotincludes 5 OFDM symbols (L_Slot=5) and each OFDM symbol is equallypartitioned into 4 RBs (N_RB_Sym=4). Accordingly, a total number of RBsin the signal frame is L_Frame*N_RB_Sym which corresponds to 80.

Numerals indicated in the signal frame of FIG. 58 refer to the order ofallocating RBs in the signal frame. Since the type1 DPs are sequentiallymapped in a frequency axis direction, it can be noted that the order ofallocating RBs is sequentially increased on the frequency axis. If theorder of allocating RBs is determined, corresponding DPs may be mappedto ultimately allocated RBs in the order of time. Assuming that anaddress to which each RB is actually mapped in the signal frame (i.e.,RB mapping address) is j, j may have a value from 0 to N_RB−1. In thiscase, if an RB input order is defined as i, i may have a value of 0, 1,2, . . . , N_RB−1 as illustrated in FIG. 58. If N_Slot=1, since the RBmapping address and the RB input order are the same (j=i), input RBs maybe sequentially mapped in ascending order of j. If N_Slot>1, RBs to bemapped to the signal frame may be partitioned and mapped according tothe number of slots, N_Slot. In this case, the RBs may be mappedaccording to a mapping rule expressed as an equation illustrated at thebottom of FIG. 58.

FIG. 59 is a view illustrating RB mapping of type2 DPs according to anembodiment of the present invention.

Like the signal frame illustrated in FIG. 58, a signal frame illustratedin FIG. 59 consists of 20 OFMD symbols (L_Frame=20) and includes 4 slots(N_Slot=4). Further, each slot includes 5 OFDM symbols (L_Slot=5) andeach OFDM symbol is equally partitioned into 4 RBs (N_RB_Sym=4).Accordingly, a total number of RBs in the signal frame isL_Frame*N_RB_Sym which corresponds to 80.

As described above in relation to FIG. 58, assuming that an address towhich each RB is actually mapped in the signal frame (i.e., RB mappingaddress) is j, j may have a value from 0 to N_RB−1. Since the type2 DPsare sequentially mapped in a time axis direction, it can be noted thatthe order of allocating RBs is sequentially increased in a time axisdirection. If the order of allocating RBs is determined, correspondingDPs may be mapped to ultimately allocated RBs in the order of time.

As described above in relation to FIG. 58, when an RB input order isdefined as i, if N_Slot=1, since j=i, input RBs may be sequentiallymapped in ascending order of j. If N_Slot>1, RBs to be mapped to thesignal frame may be partitioned and mapped according to the number ofslots, N_Slot. In this case, the RBs may be mapped according to amapping rule expressed as an equation illustrated at the bottom of FIG.59.

The equations illustrated in FIGS. 58 and 59 to express the mappingrules have no difference according to the types of DPs. However, sincethe type1 DPs are mapped in a frequency axis direction while the type2DPs are mapped in a time axis direction, different RB mapping resultsare achieved due to the difference in mapping direction.

FIG. 60 is a view illustrating RB mapping of type3 DPs according to anembodiment of the present invention.

Like the signal frames illustrated in FIGS. 23 and 24, a signal frameillustrated in FIG. 60 consists of 20 OFMD symbols (L_Frame=20) andincludes 4 slots (N_Slot=4). Further, each slot includes 5 OFDM symbols(L_Slot=5) and each OFDM symbol is equally partitioned into 4 RBs(N_RB_Sym=4). Accordingly, a total number of RBs in the signal frame isL_Frame*N_RB_Sym which corresponds to 80.

An RB mapping address of the type3 DPs may be calculated according to anequation illustrated at the bottom of FIG. 60. That is, if N_Slot=1, theRB mapping address of the type3 DPs is the same as the RB mappingaddress of the type2 DPs. The type2 and type3 DPs are the same in thatthey are sequentially mapped in a time axis direction but are differentin that the type2 DPs are mapped to the end of a first frequency of thesignal frame and then continuously mapped from a second frequency of afirst OFDM symbol while the type3 DPs are mapped to the end of a firstfrequency of a slot and then continuously mapped from a second frequencyof a first OFDM symbol of the slot in a time axis direction. Due to thisdifference, when the type3 DPs are used, time diversity may berestricted by L_Slot and power saving may be achieved on L_Slot basis.

FIG. 61 is a view illustrating RB mapping of type1 DPs according toanother embodiment of the present invention.

FIG. 61(a) illustrates an RB mapping order in a case when type1 DP0, DP1and DP2 are allocated to available RBs in a signal frame, and FIG. 61(b)illustrates an RB mapping order in a case when each of type1 DP0, DP1and DP2 is partitioned and allocated to RBs included in different slotsin a signal frame. Numerals indicated in the signal frame refer to theorder of allocating RBs. If the order of allocating RBs is determined,corresponding DPs may be mapped to ultimately allocated RBs in the orderof time.

FIG. 61(a) illustrates an RB mapping order in a case when N_Slot=1 and{DP0, DP1, DP2}={31, 15, 34}.

Specifically, DP0 may be mapped to RBs in a frequency axis directionaccording to the order of the RBs and, if an OFDM symbol is completelyfilled, move to a next OFDM symbol on the time axis to be continuouslymapped in a frequency axis direction. Accordingly, if DP0 is mapped toRB0 to RB30, DP1 may be continuously mapped to RB31 to RB45 and DP2 maybe mapped to RB46 to RB79.

To extract RBs to which a corresponding DP is mapped, the broadcastsignal reception apparatus according to an embodiment of the presentinvention needs type information of each DP (DP_Type) and the number ofequally partitioned slots (N_Slot), and needs signaling informationincluding DP start address information of each DP (DP_RB_St), FEC blocknumber information of each DP to be mapped to a signal frame(DP_N_Block), start address information of an FEC block mapped in afirst RB (DP_FEC_St), etc.

Accordingly, the broadcast signal transmission apparatus according to anembodiment of the present invention may also transmit theabove-described signaling information.

FIG. 61(b) illustrates an RB mapping order in a case when N_Slot=4 and{DP0, DP1, DP2}={31, 15, 34}.

Specifically, FIG. 61(b) shows a result of partitioning DP0, DP1 and DP2and then sequentially mapping the partitions of each DP to slots on anRB basis in the same manner as the case in which N_Slot=1. An equationexpressing a rule for partitioning RBs of each DP is illustrated at thebottom of FIG. 61. In the equation illustrated in FIG. 61, parameters s,N_RB_DP and N_RB_DP(s) may be defined as follows.

s: Slot index, s=0, 1, 2, . . . , N_Slot−1

N_RB_DP: Number of RBs of a DP to be mapped to a signal frame

N_RB_DP(s): Number of RBs of a DP to be mapped to a slot of slot index s

According to an embodiment of the present invention, since N_RB_DP=31for DP0, according to the equation illustrated in FIG. 61, the number ofRBs of DP0 to be mapped to a first slot may be N_RB_DP(0)=8, the numberof RBs of DP0 to be mapped to a second slot may be N_RB_DP(1)=8, thenumber of RBs of DP0 to be mapped to a third slot may be N_RB_DP(2)=8,and the number of RBs of DP0 to be mapped to a fourth slot may beN_RB_DP(3)=7. In the present invention, the numbers of RBs of DP0partitioned to be mapped to the slots may be expressed as {8, 8, 8, 7}.

In the same manner, DP1 may be partitioned into {4, 4, 4, 3} and DP2 maybe partitioned into {9, 9, 8, 8}.

The RBs of each partition of a DP may be sequentially mapped in eachslot using the method of the above-described case in which N_Slot=1. Inthis case, to equally fill all slots, the partitions of each DP may besequentially mapped from a slot having a smaller slot index s amongslots to which a smaller number of RBs of other DPs are allocated.

In the case of DP1, since RBs of DP0 are partitioned into {8, 8, 8, 7}and mapped to the slots in the order of s=0, 1, 2, 3, it can be notedthat the smallest number of RBs of DP0 are mapped to the slot having aslot index s=3. Accordingly, RBs of DP1 may be partitioned into {4, 4,4, 3} and mapped to the slots in the order of s=3, 0, 1, 2. In the samemanner, since the smallest number of RBs of DP0 and DP1 are allocated toslots having slot index s=2 and 3 but s=2 is smaller, RBs of DP2 may bepartitioned into {9, 9, 8, 8} and mapped to the slots in the order ofs=2, 3, 0, 1.

FIG. 62 is a view illustrating RB mapping of type1 DPs according toanother embodiment of the present invention.

FIG. 62 illustrates an embodiment in which the above-described RBmapping address of the type1 DPs is equally applied. An equationexpressing the above-described RB mapping address is illustrated at thebottom of FIG. 62. Although a mapping method and procedure in FIG. 62are different from those described above in relation to FIG. 61, sincemapping results thereof are the same, the same mapping characteristicsmay be achieved. According to the mapping method of FIG. 62, RB mappingmay be performed using a single equation irrespective of the value ofN_Slot.

FIG. 63 is a view illustrating RB mapping of type1 DPs according toanother embodiment of the present invention.

FIG. 63(a) illustrates an RB mapping order in a case when type1 DP0, DP1and DP2 are allocated to only some RBs in a signal frame, and FIG. 63(b)illustrates an RB mapping order in a case when each of type1 DP0, DP1and DP2 is partitioned and allocated to only some RBs included indifferent slots in a signal frame. Numerals indicated in the signalframe refer to the order of allocating RBs. If the order of allocatingRBs is determined, corresponding DPs may be mapped to ultimatelyallocated RBs in the order of time.

FIG. 63(a) illustrates an RB mapping order in a case when N_Slot=1 and{DP0, DP1, DP2}={7, 5, 6}.

Specifically, DP0 may be mapped to RBs in a frequency axis directionaccording to the order of the RBs and, if an OFDM symbol is completelyfilled, move to a next OFDM symbol on the time axis to be continuouslymapped in a frequency axis direction. Accordingly, if DP0 is mapped toRB0 to RB6, DP1 may be continuously mapped to RB7 to RB11 and DP2 may bemapped to RB12 to RB17.

FIG. 63(b) illustrates an RB mapping order in a case when N_Slot=4 and{DP0, DP1, DP2}={7, 5, 6}.

FIG. 63(b) illustrates embodiments in which RBs of each DP arepartitioned according to the RB partitioning rule described above inrelation to FIG. 61 and are mapped to a signal frame. Detailedprocedures thereof have been described above and thus are not describedhere.

FIG. 64 is a view illustrating RB mapping of type2 DPs according toanother embodiment of the present invention.

FIG. 64(a) illustrates an RB mapping order in a case when type2 DP0, DP1and DP2 are allocated to available RBs in a signal frame, and FIG. 64(b)illustrates an RB mapping order in a case when each of type2 DP0, DP1and DP2 is partitioned and allocated to RBs included in different slotsin a signal frame. Numerals indicated in the signal frame refer to theorder of allocating RBs. If the order of allocating RBs is determined,corresponding DPs may be mapped to ultimately allocated RBs in the orderof time.

FIG. 64(a) illustrates an RB mapping order in a case when N_Slot=1 and{DP0, DP1, DP2}={31, 15, 34}.

Since RBs of type2 DPs are mapped to the end of a first frequency of thesignal frame and then continuously mapped from a second frequency of afirst OFDM symbol, time diversity may be achieved. Accordingly, if DP0is mapped to RB0 to RB19 on a time axis and then continuously mapped toRB20 to RB30 of the second frequency, DP1 may be mapped to RB31 to RB45in the same manner and DP2 may be mapped to RB46 to RB79.

To extract RBs to which a corresponding DP is mapped, the broadcastsignal reception apparatus according to an embodiment of the presentinvention needs type information of each DP (DP_Type) and the number ofequally partitioned slots (N_Slot), and needs signaling informationincluding DP start address information of each DP (DP_RB_St), FEC blocknumber information of each DP to be mapped to a signal frame(DP_N_Block), start address information of an FEC block mapped in afirst RB (DP_FEC_St), etc.

Accordingly, the broadcast signal transmission apparatus according to anembodiment of the present invention may also transmit theabove-described signaling information.

FIG. 64(b) illustrates an RB mapping order in a case when N_Slot=4 and{DP0, DP1, DP2}={31, 15, 34}.

A first signal frame of FIG. 64(b) shows a result of performing RBmapping according to the RB partitioning rule described above inrelation to FIG. 61, and a second signal frame of FIG. 64(b) shows aresult of performing RB mapping by equally applying the above-describedRB mapping address of the type2 DPs. Although mapping methods andprocedures of the above two cases are different, since mapping resultsthereof are the same, the same mapping characteristics may be achieved.In this case, RB mapping may be performed using a single equationirrespective of the value of N_Slot.

FIG. 65 is a view illustrating RB mapping of type2 DPs according toanother embodiment of the present invention.

FIG. 65(a) illustrates an RB mapping order in a case when type2 DP0, DP1and DP2 are allocated to only some RBs in a signal frame, and FIG. 65(b)illustrates an RB mapping order in a case when each of type2 DP0, DP1and DP2 is partitioned and allocated to only some RBs included indifferent slots in a signal. frame. Numerals indicated in the signalframe refer to the order of allocating RBs. If the order of allocatingRBs is determined, corresponding DPs may be mapped to ultimatelyallocated RBs in the order of time.

FIG. 65(a) illustrates an RB mapping order in a case when N_Slot=1 and{DP0, DP1, DP2}={7, 5, 6}.

Specifically, DP0 may be mapped to RBs in a time axis directionaccording to the order of the RBs and, if DP0 is mapped to RB0 to RB6,DP1 may be continuously mapped to RB7 to RB11 and DP2 may be mapped toRB12 to RB17.

FIG. 65(b) illustrates an RB mapping order in a case when N_Slot=4 and{DP0, DP1, DP2}={7, 5, 6}.

FIG. 65(b) illustrates embodiments in which RBs of each DP arepartitioned according to the RB partitioning rule described above inrelation to FIG. 61 and are mapped to a signal frame. Detailedprocedures thereof have been described above and thus are not describedhere.

FIG. 66 is a view illustrating RB mapping of type3 DPs according toanother embodiment of the present invention.

FIG. 66(a) illustrates an RB mapping order in a case when each of type3DP0, DP1 and DP2 is partitioned and allocated to RBs included indifferent slots in a signal frame, and FIG. 66(b) illustrates an RBmapping order in a case when each of type3 DP0, DP1 and DP2 ispartitioned and allocated to only some RBs included in a slot in asignal frame. Numerals indicated in the signal frame refer to the orderof allocating. RBs. If the order of allocating RBs is determined,corresponding DPs may be mapped to ultimately allocated RBs in the orderof time.

FIG. 66(a) illustrates an RB mapping order in a case when N_Slot=4 and{DP0, DP1, DP2}={31, 15, 34}.

A first signal frame of FIG. 66(a) illustrates an embodiment in whichthe above-described RB mapping address of the type3 DPs is equallyapplied. A second signal frame of FIG. 66(a) illustrates an embodimentin which, when the number of RBs of a DP is greater than that of a slot,time diversity is achieved by changing a slot allocation order.Specifically, the second signal frame of FIG. 66(a) corresponds to anembodiment in which, when the number of RBs of DP0 allocated to a firstslot of the first signal frame is greater than that of the first slot,the remaining RBs of DP0 are allocated to a third slot.

FIG. 66(b) illustrates an RB mapping order in a case when N_Slot=4 and{DP0, DP1, DP2}={7, 5, 6}.

Further, to extract RBs to which a corresponding DP is mapped, thebroadcast signal reception apparatus according to an embodiment of thepresent invention needs type information of each DP (DP_Type) and thenumber of equally partitioned slots (N_Slot), and needs signalinginformation including DP start address information of each DP(DP_RB_St), FEC block number information of each DP to be mapped to asignal frame (DP_N_Block), start address information of an FEC blockmapped in a first RB (DP_FEC_St), etc.

Accordingly, the broadcast signal transmission apparatus according to anembodiment of the present invention may also transmit theabove-described signaling information.

FIG. 67 is a view illustrating RB mapping of type3 DPs according toanother embodiment of the present invention.

FIG. 67 illustrates RB mapping in a case when N_Slot=1 and {DP0, DP1,DP2}={7, 5, 6}. As illustrated in FIG. 67, RBs of each DP may be mappedon an arbitrary block basis in a signal frame. In this case, thebroadcast signal reception apparatus according to an embodiment of thepresent invention needs additional signaling information as well as theabove-described signaling information to extract RBs to which acorresponding DP is mapped.

As such, the present invention may exemplarily describe a case in whichDP end address information of each DP (DP_RB_Ed) is additionallytransmitted. Accordingly, the broadcast signal transmission apparatusaccording to an embodiment of the present invention may map RBs of theDP on an arbitrary block basis and transmit the above-describedsignaling information, and the broadcast signal reception apparatusaccording to an embodiment of the present invention may detect anddecode the RBs of the DP mapped on an arbitrary block basis, usingDP_RB_St information and DP_RB_Ed information included in theabove-described signaling information. When this method is used, free RBmapping is enabled and thus DPs may be mapped with different RB mappingcharacteristics.

Specifically, as illustrated in FIG. 67, RBs of DP0 may be mapped in acorresponding block in a time axis direction to achieve time diversitylike type2 DPs, RBs of DP1 may be mapped in a corresponding block in afrequency axis direction to achieve the power saving effect like type1DPs. Besides, RBs of DP2 may be mapped in a corresponding block inconsideration of time diversity and power saving like type3 DPs.

Further, even in a case when RBs are not mapped in the wholecorresponding block like DP1, the broadcast signal reception apparatusmay accurately detect the locations of RBs to be acquired, using theabove-described signaling information, e.g., DP_FEC_St information,DP_N_Block information, DP_RB_St information and DP_RB_Ed information,and thus a broadcast signal may be efficiently transmitted and received.

FIG. 68 is a view illustrating signaling information according to anembodiment of the present invention.

FIG. 68 illustrates the above-described signaling information related toRB mapping according to DP types, and the signaling information may betransmitted using signaling through a PLS (hereinafter referred to asPLS signaling) or in-band signaling.

Specifically, FIG. 68(a) illustrates signaling information transmittedthrough a PLS, and FIG. 68(b) illustrates signaling informationtransmitted through in-band signaling.

As illustrated in FIG. 68, the signaling information related to RBmapping according to DP types may include N_Slot information, DP_Typeinformation, DP_N_Block information, DP_RB_St information, DP_FEC_Stinformation and DP_N_Block information.

The signaling information transmitted through PLS signaling is the sameas the signaling information transmitted through in-band signaling.However, a PLS includes information about all DPs included in acorresponding signal frame for service acquisition and thus thesignaling information other than N_Slot information and DP_Typeinformation may be defined within a DP loop for defining informationabout every DP. On the other hand, in-band signaling is used to acquirea corresponding DP and thus is transmitted for each DP. As such, in-bandsignaling is different from PLS signaling in that a DP loop for defininginformation about every DP is not necessary. A brief description is nowgiven of the signaling information.

N_Slot information: Information indicating the number of slotspartitioned form a signal frame, which may have the size of 2 bits.According to an embodiment of the present invention, the number of slotsmay be 1, 2, 4, 8.

DP_Type information: Information indicating the type of a DP, which maybe one of type 1, type 2 and type 3 as described above. This informationis extensible according to the intention of a designer and may have thesize of 3 bits.

DP_N_Block_Max information: Information indicating the maximum number ofFEC blocks of a corresponding DP or a value equivalent thereto, whichmay have a size of 10 bits.

DP_RB_St information: Information indicating an address of a first RB ofa corresponding DP, and the address of an RB may be expressed on an RBbasis. This information may have a size of 8 bits.

DP_FEC_St information: Information indicating a first address of an FECblock of a corresponding DP to be mapped to a signal frame, and theaddress of an FEC block may be expressed on a cell basis. Thisinformation may have a size of 13 bits.

DP_N_Block information: Information indicating the number of FEC blocksof a corresponding DP to be mapped to a signal frame or a valueequivalent thereto, which may have a size of 10 bits.

The above-described signaling information may vary name, size, etc.thereof according to the intention of a designer in consideration of thelength of a signal frame, the size of time interleaving, the size of RB,etc.

Since PLS signaling and in-band signaling have a difference according touses thereof as described above, for more efficient transmission,signaling information may be omitted for PLS signaling and in-bandsignaling as described below.

First, a PLS includes information about all DPs included in acorresponding signal frame. Accordingly, DPs are completely andsequentially mapped to the signal frame in the order of DP0, DP1, DP2, .. . , the broadcast signal reception apparatus may perform calculationto achieve DP_RB_St information. In this case, DP_RB_St information maybe omitted.

Second, in the case of in-band signaling, the broadcast signal receptionapparatus may acquire DP_FEC_St information of a next signal frame usingDP_N_Block information of a corresponding DP. Accordingly, DP_FEC_Stinformation may be omitted.

Third, in the case of in-band signaling, when N_Slot information,DP_Type information and DP_N_Block_Max information which influencemapping of a corresponding DP are changed, a 1-bit signal indicatingwhether the corresponding information is changed may be used, or thechange may be signaled. In this case, additional N_Slot information,DP_Type information and DP_N_Block_Max information may be omitted.

That is, DP_RB_St information may be omitted in the PLS, and signalinginformation other than DP_RB_St information and DP_N_Block informationmay be omitted in in-band signaling. This is variable according to theintention of a designer.

FIG. 69 is a graph showing the number of bits of a PLS according to thenumber of DPs according to an embodiment of the present invention.

Specifically, FIG. 69 shows an increase in number of bits for PLSsignaling in a case when signaling information related to RB mappingaccording to DP types is transmitted through a PLS, as the number of DPsis increased.

A dashed line refers to a case in which every related signalinginformation is transmitted (Default signaling), and a solid line refersto a case in which the above-described types of signaling informationare omitted (Efficient signaling). As the number of DPs is increased, ifcertain types of signaling information are omitted, it is noted that thenumber of saved bits is linearly increased.

FIG. 70 is a view illustrating a procedure for demapping DPs accordingto an embodiment of the present invention.

As illustrated in the top of FIG. 70, the broadcast signal transmissionapparatus according to an embodiment of the present invention maytransmit contiguous signal frames 35000 and 35100. The configuration ofeach signal frame is as described above.

As described above, when the broadcast signal transmission apparatusmaps DPs of different types to a corresponding signal frame on an RBbasis and transmits the signal frame, the broadcast signal receptionapparatus may acquire a corresponding DP using the above-describedsignaling information related to RB mapping according to DP types.

As described above, the signaling information related to RB mappingaccording to DP types may be transmitted through a PLS 35010 of thesignal frame or through in-band signal 35020. FIG. 70(a) illustratessignaling information related to RB mapping according to DP types, whichis transmitted through the PLS 35010, and FIG. 70(b) illustratessignaling information related to RB mapping according to DP types, whichis transmitted through in-band signaling 35020. In-band signaling 35020is processed, e.g., coded, modulated, and time-interleaved, togetherwith data included in the corresponding DP, and thus may be indicated asbeing included as parts of data symbols in the signal frame. Each typeof signaling information has been described above and thus is notdescribed here.

As illustrated in FIG. 70, the broadcast signal reception apparatus mayacquire the signaling information related to RB mapping according to DPtypes, which is included in the PLS 35010, and thus may demap andacquire DPs mapped to the corresponding signal frame 35000. Further, thebroadcast signal reception apparatus may acquire the signalinginformation related to RB mapping according to DP types, which istransmitted through in-band signaling 35020, and thus may demap DPsmapped to the next signal frame 35100.

PLS Protection&Structure (Repetition)

FIG. 71 is a view illustrating exemplary structures of three types ofmother codes applicable to perform LDPC encoding on PLS data in an FECencoder module according to another embodiment of the present invention.

PLS-pre data and PLS-post data output from the above-described PLSgeneration module 4300 are independently input to the BB scramblermodule 4400. In the following description, the PLS-pre data and thePLS-post data may be collectively called PLS data. The BB scramblermodule 4400 may perform initialization to randomize the input PLS data.The BB scrambler module 4400 may initialize the PLS data located and tobe transmitted in frame, on a frame basis.

If the PLS located and to be transmitted in frame includes informationabout a plurality of frames, the BB scrambler module 4400 may initializethe PLS data on a frame basis. An example thereof is the case of a PLSrepetition frame structure to be described below. According to anembodiment of the present invention, PLS repetition refers to a frameconfiguration scheme for transmitting PLS data for a current frame andPLS data for a next frame together in the current frame. When PLSrepetition is applied, the BB scrambler module 4400 may independentlyinitialize the PLS data for the current frame and the PLS data for thenext frame. A detailed description of PLS repetition will be givenbelow.

The BB scrambler module 4400 may randomize the PLS-pre data and thePLS-post data initialized on a frame basis.

The randomized PLS-pre data and the PLS-post data are input to thecoding & modulation module 5300. The randomized PLS-pre data and therandomized PLS-post data may be respectively input to the FEC encodermodules 5310 included in the coding & modulation module 5300. The FECencoder modules 5310 may respectively perform BCH encoding and LDPCencoding on the input PLS-pre data and the PLS-post data. Accordingly,the FEC encoder modules 5310 may respectively perform LDPC encoding onthe randomized PLS-pre data and the randomized PLS-post data input tothe FEC encoder modules 5310.

BCH parity may be added to the randomized PLS data input to the FECencoder modules 5310 due to BCH encoding, and then LDPC encoding may beperformed on the BCH-encoded data. LDPC encoding may be performed basedon one of mother code types having different sizes in informationportion (hereinafter, the size of information portion is called K_ldpc)according to the size of input data including BCH parity (hereinafter,the size of data input to an LDPC encoder module is called N_BCH). TheFEC encoder module 5310 may shorten data of an information portion of anLDPC mother code corresponding to the difference 36010 in size betweenK_ldpc and N_BCH, to 0 or 1, and may puncture a part of data included ina parity portion, thereby outputting a shortened/punctured LDPC code.The LDPC encoder module may perform LDPC encoding on the input PLS dataor the BCH-encoded PLS data based on the shortened/punctured LDPC codeand output the LDPC-encoded PLS data.

Here, BCH encoding is omittable according to the intention of adesigner. If BCH encoding is omitted, the FEC encoder module 5310 maygenerate an LDPC mother code by encoding the PLS data input to the FECencoder module 5310. The FEC encoder module 5310 may shorten data of aninformation portion of the generated LDPC mother code corresponding tothe difference 36010 in size between K_ldpc and PLS data, to 0 or 1, andmay puncture a part of data included in a parity portion, therebyoutputting a shortened/punctured LDPC code. The FEC encoder module 5310may perform LDPC encoding on the input PLS data based on theshortened/punctured LDPC code and output the LDPC-encoded PLS data.

FIG. 71(a) illustrates an exemplary structure of mother code type1.Here, mother code type1 has a code rate of ⅙. FIG. 71(b) illustrates anexemplary structure of mother code type2. Here, mother code type2 has acode rate of ¼. FIG. 71(c) illustrates an exemplary structure of mothercode type3. Here, mother code type3 has a code rate of ⅓.

As illustrated in FIG. 71, each mother code may include an informationportion and a parity portion. According to an embodiment of the presentinvention, the size of data corresponding to an information portion 3600of a mother code may be defined as K_ldpc. K_ldpc of mother code type1,mother code type2 and mother code type3 may be respectively calledk_ldpc1, k_ldpc2 and k_ldpc3.

A description is now given of an LDPC encoding procedure performed by anFEC encoder module based on mother code type1 illustrated in FIG. 71(a).In the following description, encoding may refer to LDPC encoding.

When BCH encoding is applied, the information portion of the mother codemay include BCH-encoded PLS data including BCH parity bits and input tothe LDPC encoder module of the FEC encoder module.

When BCH encoding is not applied, the information portion of the mothercode may include PLS data input to the LDPC encoder module of the FECencoder module.

The size of the PLS data input to the FEC encoder module may varyaccording to the size of additional information (management information)to be transmitted and the size of data of transmission parameters. TheFEC encoder module may insert “0” bits to the BCH-encoded PLS data. IfBCH encoding is not performed, the FEC encoder module may insert “0”bits to the PLS data.

The present invention may provide three types of dedicated mother codesused to perform the above-described LDPC encoding according to anotherembodiment. The FEC encoder module may select a mother code according tothe size of PLS data, and the mother code selected by the FEC encodermodule according to the size of PLS data may be called a dedicatedmother code. The FEC encoder module may perform LDPC encoding based onthe selected dedicated mother code.

According to an embodiment of the present invention, the size 36000 ofK_ldpc1 of mother code type1 may be assumed as ½ of the size of K_ldpc2of mother code type2 and ¼ of the size of K_ldpc3 of mother code type3.The relationship among the sizes of K_ldpc of mother code types isvariable according to the intention of a designer. The designer maydesign a mother code having a small size of K_ldpc to have a low coderate. To maintain a constant signaling protection level of PLS datahaving various sizes, an effective code rate after shortening andpuncturing should be lowered as the size of PLS data is small. To reducethe effective code rate, a parity ratio of a mother code having a smallsize of K_ldpc may be increased.

If the PLS data has an excessively large size and thus cannot be encodedbased on one of a plurality of mother code types by the FEC encodermodule, the PLS data may be split into a plurality of pieces forencoding. Here, each piece of the PLS data may be called fragmented PLSdata. The above-described procedure for encoding the PLS data by the FECencoder module may be replaced with a procedure for encoding eachfragmented PLS data if the PLS data has an excessively large size andthus cannot be encoded based on one of a plurality of mother code typesby the FEC encoder module.

When the FEC encoder module encodes mother code type1, to secure asignaling protection level in a very low signal to noise ratio (SNR)environment, payload splitting may be performed. The length of parity ofmother code type1 may be increased due to a portion 36020 for executinga payload splitting mode. A detailed description of the mother codeselection method and the payload splitting mode will be given below.

If the FEC encoder module encodes PLS data having various sizes based ona single mother code type having a large size of K_ldpc, a coding gainmay be rapidly reduced. For example, when the above-described FECencoder module performs shortening using a method for determining ashortening data portion (e.g., K_ldpc−N_BCH), since K_ldpc is constant,small-sized PLS data is shortened more than large-sized PLS data.

To solve the above-described problem, the FEC encoder module accordingto an embodiment of the present invention may apply a mother code typecapable of achieving an optimal coding gain among a plurality of mothercode types differently according to the size of PLS data.

The FEC encoder module according to an embodiment of the presentinvention may restrict the size of a portion to be shortened by the FECencoder module to achieve an optimal coding gain. Since the FEC encodermodule restricts the size 36010 of a shortening portion to be shortenedto a certain ratio of K_ldpc 36000 of each mother code, a coding gain ofa dedicated mother code of each PLS data may be constantly maintained.The current embodiment shows an example in which shortening can beperformed up to 50% of the size of K_ldpc. Accordingly, when theabove-described FEC encoder module determines a shortening data portionas the difference between K_ldpc and N_BCH, if the difference betweenK_ldpc and N_BCH is greater than ½ of K_ldpc, the FEC encoder module maydetermine the size of a data portion to be shortened by the FEC encodermodule as K_ldpc*½ instead of K_ldpc-N_BCH.

LDPC encoding procedures performed by the FEC encoder module based onmother code type2 and mother code type3 illustrated in FIGS. 36(b) and36(c) may be performed in the same manner as the above-described LDPCencoding procedure performed by the FEC encoder module based on mothercode type1 illustrated in FIG. 71(a).

The FEC encoder module may perform encoding based on an extended LDPCcode by achieving an optimal coding gain by encoding PLS data havingvarious sizes based on a single mother code.

However, a coding gain achievable when encoding is performed based on anextended LDPC code is approximately 0.5 dB lower than the coding gainachievable when encoding is performed based on dedicated mother codesoptimized to different sizes of PLS data as described above. Thus, ifthe FEC encoder module according to an embodiment of the presentinvention encodes PLS data by selecting a mother code type structureaccording to the size of PLS data, redundancy data may be reduced andPLS signaling protection capable of ensuring the same receptionperformance may be designed.

FIG. 72 is a flowchart of a procedure for selecting a mother code typeused for LDPC encoding and determining the size of shortening accordingto another embodiment of the present invention.

A description is now given of a procedure for selecting a mother codetype according to the size of PLS data (payload size) to be LDPC-encodedand determining the size of shortening by the FEC encoder module. Thefollowing description is assumed that all operations below are performedby the FEC encoder module.

It is checked whether an LDPC encoding mode is a normal mode or apayload splitting mode (S37000). If the LDPC encoding mode is a payloadsplitting mode, mother code1 may be selected irrespective of the size ofPLS data and the size of shortening is determined based on the size ofK_ldpc of mother code type1 (k_ldpc1) (S37060). A detailed descriptionof the payload splitting mode will be given below.

If the LDPC encoding mode is a normal mode, the FEC encoder moduleselects a mother code type according to the size of PLS data. Adescription is now given of the procedure for selecting a mother codetype in the normal mode by the FEC encoder module.

Num_ldpc refers to the number of fragmented PLS data which can beincluded in a single piece of PLS data. Isize_ldpc refers to the size offragmented PLS data input to the FEC encoder module. Num_ldpc3 may bedetermined as a rounded-up value of a value obtained by dividing thesize of input PLS data (payload size) by k_ldpc3 for encoding. The valueof isize_ldpc3 may be determined as a rounded-up value of a valueobtained by dividing the size of PLS data (payload size) by thedetermined num_ldpc3 (S37010). It is determined whether the value ofisize_ldpc3 is in a range greater than k_ldpc2 and equal to or less thank_ldpc3 (S37020). If the size of isize_ldpc3 is in a range greater thank_ldpc2 and equal to or less than k_ldpc3, mother code type3 isdetermined. In this case, the size of shortening may be determined basedon a difference value between k_ldpc3 and isize_ldpc3 (S37021).

If the value of isize_ldpc3 is not in a range greater than k_ldpc2 andequal to or less than k_ldpc3, a rounded-up value of a value obtained bydividing the size of PLS data (marked as “payload size” in FIG. 72) byk_ldpc2 is determined as num_ldpc2. The value of isize_ldpc2 may bedetermined as a rounded-up value of a value obtained by dividing thesize of PLS data (payload size) by the determined num_ldpc2 (S37030). Itis determined whether the value of isize_ldpc2 is in a range greaterthan k_ldpc1 and equal to or less than k_ldpc2 (S37040). If the value ofisize_ldpc2 is in a range greater than k_ldpc1 and equal to or less thank_ldpc2, mother code type2 is determined. In this case, the size ofshortening may be determined based on a difference value between k_ldpc2and isize_ldpc2 (S37041).

If the value of isize_ldpc2 is in not a range greater than k_ldpc1 andequal to or less than k_ldpc2, a rounded-up value of a value obtained bydividing the size of PLS data (payload size) by k_ldpc1 is determined asnum_ldpc1. The value of isize_ldpc1 may be determined as a rounded-upvalue of a value obtained by dividing the size of PLS data (payloadsize) by the determined num_ldpc1 (S37050). In this case, mother codetype1 is determined and the size of shortening may be determined basedon a difference value between k_ldpc1 and isize_ldpc1 (S37060).

The above-described num_ldpc and isize_ldpc may have different valuesaccording to the size of PLS data. However, k_ldpc1, k_ldpc2 and k_ldpc3according to the mother code type are not influenced by the size of PLSdata and have constant values.

FIG. 73 is a view illustrating a procedure for encoding adaptationparity according to another embodiment of the present invention.

FIG. 73(a) illustrates an example of PLS data input to the FEC encodermodule for LDPC encoding.

FIG. 73(b) illustrates an exemplary structure of an LDPC code afterperforming LDPC encoding and before performing shortening andpuncturing.

FIG. 73(c) illustrates an exemplary structure of an LDPC code afterperforming LDPC encoding, shortening and puncturing (38010) (hereinafterreferred to as a shortened/punctured LDPC code), which is output fromthe FEC encoder module.

FIG. 73(d) illustrates an exemplary structure of a code output by addingadaptation parity (38011) to the LDPC code which is LDPC-encoded,shortened and punctured by the FEC encoder module, according to anotherembodiment of the present invention. Here, a scheme for outputting thecode by adding adaptation parity (38011) to the shortened/punctured LDPCcode by the FEC encoder module is called an adaptation parity scheme.

To maintain a signaling protection level, the FEC encoder module mayperform LDPC-encode and then shorten the PLS data, puncture (38010) someof parity bits, and thus output the shortened/punctured LDPC code. In apoor reception environment, the signaling protection level needs to bestrengthened compared to the robustness constantly supported by abroadcast system, i.e., a constant target threshold of visibility (TOV).According to an embodiment of the present invention, to strengthen thesignaling protection level, an LDPC code may be output by addingadaptation parity bits to the shortened/punctured LDPC code. Theadaptation parity bits may be determined as some parity bits (38011) ofthe parity bits (38010) punctured after LDPC encoding.

FIG. 73(c) illustrates a basic target TOV in a case when an effectivecode rate is approximately ⅓. According to an embodiment of the presentinvention, if the FEC encoder module adds the adaptation parity bits(38011), actually punctured parity bits may be reduced. The FEC encodermodule may adjust the effective code rate to approximately ¼ by addingadaptation parity bits as illustrated in FIG. 73(d). According to anembodiment of the present invention, a mother code used for LDPCencoding may additionally include a certain number of parity bits toacquire the adaptation parity bits 38011. Accordingly, the coding rateof a mother code used for adaptation parity encoding may be designed tobe lower than the code rate of an original mother code.

The FEC encoder module may output the added parity (38011) included inthe LDPC code by arbitrarily reducing the number of punctured paritybits. A diversity gain may be achieved by including the output addedparity (38011) included in the LDPC code, in a temporally previous frameand transmitting the previous frame via a transmitter. The end of aninformation portion of a mother code is shortened and the end of aparity portion of the mother code is punctured in FIG. 73(b). However,this merely corresponds to an exemplary embodiment and the shorteningand puncturing portions in the mother code may vary according to theintention of a designer.

FIG. 74 is a view illustrating a payload splitting mode for splittingPLS data input to the FEC encoder module before LDPC-encoding the inputPLS data according to another embodiment of the present invention. Inthe following description, the PLS data input to the FEC encoder modulemay be called payload.

FIG. 74(a) illustrates an example of PLS data input to the FEC encodermodule for LDPC encoding.

FIG. 74(b) illustrates an exemplary structure of an LDPC code obtainedby LDPC-encoding each split piece of payload. The structure of the LDPCcode illustrated in FIG. 74(b) is the structure before performingshortening/puncturing.

FIG. 74(c) illustrates an exemplary structure of a shortened/puncturedLDPC code output from the FEC encoder module according to anotherembodiment of the present invention. The structure of theshortened/punctured LDPC code illustrated in FIG. 74(c) is the structureof the shortened/punctured LDPC code output when a payload splittingmode is applied to the FEC encoder module.

Payload splitting is performed by the FEC encoder module to achieve therobustness strengthened compared to a constant target TOV for signaling.

As illustrated in FIG. 74(b), the payload splitting mode is a mode forsplitting PLS data before LDPC encoding and performing LDPC encoding oneach split piece of the PLS data by the FEC encoder module.

As illustrated in FIG. 74(c), in the payload splitting mode, the inputPLS data may be encoded and shortened/punctured using only a mother codetype having the lowest code rate among mother code types provided by theFEC encoder module (e.g., mother code type1 according to the currentembodiment).

A method for selecting one of three mother code types based on the sizeof PLS data and performing LDPC encoding on the LDPC encoding based onthe selected mother code type to adjust a signaling protection level byFEC encoder module has been described above. However, if a mother codetype having the highest code rate is selected among mother code typesprovided by the FEC encoder module (e.g., mother code type3 according tothe current embodiment), the signaling protection level may berestricted. In this case, the FEC encoder module may apply the payloadsplitting mode to the PLS data and LDPC-encode every piece of the PLSdata using only a mother code type having the lowest code rate amongmother code types provided by the FEC encoder module, thereby adjustingthe signaling protection level to be low. When the payload splittingmode is used, the FEC encoder module may adjust the size of punctureddata according to a strengthened target TOV after shortening.

According to the previous embodiment of the present invention, when theFEC encoder module does not use the payload splitting mode for LDPCencoding, the effective code rate of the shortened/punctured LDPC codewas approximately ⅓. However, in FIG. 74(c), the effective code rate ofthe output LDPC code to which the payload splitting mode is applied bythe FEC encoder module is approximately 11/60. Accordingly, theeffective code rate of the output LDPC code to which the payloadsplitting mode is applied may be reduced.

The end of an information portion of an LDPC code is shortened and theend of a parity portion of the LDPC code is punctured in FIG. 74(b).However, this merely corresponds to an exemplary embodiment and theshortening and puncturing portions in the LDPC code may vary accordingto the intention of a designer.

FIG. 75 is a view illustrating a procedure for performing PLS repetitionand outputting a frame by the frame structure module 1200 according toanother embodiment of the present invention.

According to another embodiment of the present invention, PLS repetitionperformed by the frame structure module corresponds to a frame structurescheme for including two or more pieces of PLS data includinginformation about two or more frames in a single frame.

A description is now given of PLS repetition according to an embodimentof the present invention.

FIG. 75(a) illustrates an exemplary structure of a plurality of piecesof PLS data encoded by the FEC encoder module.

FIG. 75(b) illustrates an exemplary structure of a frame including aplurality of pieces of encoded PLS data due to PLS repetition by theframe structure module.

FIG. 75(c) illustrates an exemplary structure of a current frameincluding PLS data of the current frame and PLS data of a next frame.

Specifically, FIG. 75(c) illustrates an exemplary structure of an nthframe (current frame) including PLS data (PLS n) of the nth frame andPLS data 40000 of an (n+1)th frame (next frame), and the (n+1)th frame(current frame) including PLS data (PLS n+1) of the (n+1)th frame andPLS data of an (n+2)th frame (next frame). A detailed description is nowgiven of FIG. 75.

FIG. 75(a) illustrates the structure in which PLS n for the nth frame,PLS n+1 for the (n+1)th frame, and PLS n+2 for the (n+2)th frame areencoded. The FEC encoder module according to another embodiment of thepresent invention may output an LDPC code by encoding static PLSsignaling data and dynamic PLS signaling data together. PLS n includingphysical signaling data of the nth frame may include static PLSsignaling data (marked as “stat”), dynamic PLS signaling data (marked as“dyn”), and parity data (marked as “parity”). Likewise, each of PLS n+1and PLS n+2 including physical signaling data of the (n+1)th frame andthe (n+2)th frame may include static PLS signaling data (marked as“stat”), dynamic PLS signaling data (marked as “dyn”), and parity data(marked as “parity”). In FIG. 75(a), l includes static PLS signalingdata and dynamic PLS signaling data, and P includes parity data.

FIG. 75(b) illustrates an example of PLS formatting for splitting thedata illustrated in FIG. 75(a) to locate the data in frames.

If PLS data transmitted by a transmitter is split according to whetherthe PLS data is changed for each frame and then transmitted by excludingredundancy data which is not changed in every frame, a receiver may havea higher PLS decoding performance. Accordingly, PLS n and PLS n+1 aremapped to the nth frame using PLS repetition, the frame structure moduleaccording to an embodiment of the present invention may split PLS n+1 toinclude the dynamic PLS signaling data of PLS n+1 and the parity data ofPLS n+1 excluding the static PLS signaling data of PLS n+1 which isrepeated from the static PLS signaling data of PLS n. A splitting schemefor transmitting PLS data of a next frame in a current frame by theframe structure module may be called PLS formatting.

Here, when the frame structure module splits PLS n+1 to be mapped to thenth frame, the parity data of PLS n+1 may be determined as a part ofparity data (marked as “P”) illustrated in FIG. 75(a), and the sizethereof can scalably vary. Parity bits of PLS data of a next frame to betransmitted in a current frame, which are determined by the framestructure module due to PLS formatting, may be called scalable parity.

FIG. 75(c) illustrates an example in which data split in FIG. 75(b) islocated in the nth frame and the (n+1)th frame.

Each frame may include a preamble, PLS-pre, PLS and service data (markedas “Data n”). A description is now given of the detailed stricture ofeach frame illustrated in FIG. 75(c). The nth frame illustrated in FIG.75(c) may include a preamble, PLS-pre, encoded PLS n, a part of encodedPLS n+1 40000, and service data (marked as “Data n”). Likewise, the(n+1)th frame may include a preamble, PLS-pre, encoded PLS n+1 40010, apart of encoded PLS n+2, and service data (marked as “Data n+1”). In thefollowing description according to an embodiment of the presentinvention, a preamble may include PLS-pre.

PLS n+1 included in the nth frame is different from that included in the(n+1)th frame in FIG. 75(c). PLS n+1 40000 included in the nth frame issplit due to PLS formatting and does not include static PLS signalingdata while PLS n+1 40010 includes static PLS signaling data.

When scalable parity is determined, the frame structure module maymaintain the robustness of PLS n+1 40000 included in the nth frame insuch a manner that a receiver can decode PLS n+1 included in the nthframe before receiving the (n+1)th frame and may consider a diversitygain achievable when PLS n+1 40000 included in the nth frame and PLS n+140010 included in the (n+1)th frame are decoded in the (n+1)th frame.

If parity bits of PLS n+1 40000 included in the nth frame are increased,data (Data n+1) included in the (n+1)th frame may be rapidly decodedbased on data achieved by decoding PLS n+1 40000 included in the nthframe before the (n+1)th frame is received. On the other hand, scalableparity included in PLS n+1 40000 may be increased and thus datatransmission may be inefficient. Further, if small scalable parity ofPLS n+1 40000 is transmitted in the n frame to achieve a diversity gainfor decoding PLS n+1 40010 included in the (n+1)th frame, the effect ofrapidly decoding service data (Dana n+1) included in the (n+1)th frameby previously decoding PLS n+1 40000 included in the n frame before the(n+1)th frame is received may be reduced.

To achieve an improved diversity gain by a receiver, the frame structuremodule according to an embodiment of the present invention may determinethe configuration of parity of PLS n+1 40000 included in the nth frameto be different from that of parity of PLS n+1 40010 included in the(n+1)th frame as much as possible in the PLS formatting procedure.

For example, if parity P of PLS n+1 includes 5 bits, the frame structuremodule may determine scalable parity of PLS n+1 which can be included inthe nth frame as second and fourth bits and determine scalable parity ofPLS n+1 which can be included in the (n+1)th frame as first, third andfifth bits. As such, if the frame structure module determines scalableparity bits not to overlap, a coding gain as well as a diversity gainmay be achieved. According to another embodiment of the presentinvention, when the frame structure module performs PLS formatting, adiversity gain of a receiver may be maximized by soft-combiningrepeatedly transmitted information before LDPC decoding.

The frame structure illustrated in FIG. 75 is merely an exemplaryembodiment of the present invention and may vary according to theintention of a designer. The order of PLS n and PLS n+1 40000 in the nthframe merely an example and PLS n+1 40000 may be located prior to PLS naccording to the intention of a designer. This may be equally applied tothe (n+1)th frame.

FIG. 76 is a view illustrating signal frame structures according toanother embodiment of the present invention.

Each of signal frames 41010 and 41020 illustrated in FIG. 76(a) mayinclude a preamble P, head/tail edge symbols E_(H)/E_(T), one or morePLS symbols PLS and a plurality of data symbols (marked as “DATA FrameN” and “DATA Frame N+1”). This is variable according to the intention ofa designer. “T_Sync” marked in each signal frame of FIGS. 41(a) and41(b) refers to a time necessary to achieve stable synchronization forPLS decoding based on information acquired from a preamble by areceiver. A description is now given of a method for allocating a PLSoffset portion by the frame structure module to ensure T_Sync time.

The preamble is located at the very front of each signal frame and maytransmit a basic transmission parameter for identifying a broadcastsystem and the type of signal frame, information for synchronization,information about modulation and coding of a signal included in theframe, etc. The basic transmission parameter may include FFT size, guardinterval information, pilot pattern information, etc. The informationfor synchronization may include carrier and phase, symbol timing andframe information. Accordingly, a broadcast signal reception apparatusaccording to another embodiment of the present invention may initiallydetect the preamble of the signal frame, identify the broadcast systemand the frame type, and selectively receive and decode a broadcastsignal corresponding to a receiver type.

Further, the receiver may acquire system information using informationof the detected and decoded preamble, and may acquire information forPLS decoding by additionally performing a synchronization procedure. Thereceiver may perform PLS decoding based on the information acquired bydecoding the preamble.

To perform the above-described function of the preamble, the preamblemay be transmitted with a robustness several dB higher than that ofservice data. Further, the preamble should be detected and decoded priorto the synchronization procedure.

FIG. 76(a) illustrates the structure of signal frames in which PLSsymbols are mapped subsequently to the preamble symbol or the edgesymbol E_(H). Since the receiver completes synchronization after a timecorresponding to T_Sync, the receiver may not decode the PLS symbolsimmediately after the PLS symbols are received. In this case, a time forreceiving one or more signal frames may be delays until the receiverdecodes the received PLS data. Although a buffer may be used for a casein which synchronization is not completed before PLS symbols of a signalframe are received, a problem in which a plurality of buffers arenecessary may be caused.

Each of signal frames 41030 and 41040 illustrated in FIG. 76(b) may alsoinclude the symbols P, E_(H), E_(T), PLS and DATA Frame N illustrated inFIG. 76(a).

The frame structure module according to another embodiment of thepresent invention may configure a PLS offset portion 41031 or 41042between the head edge symbol E_(H) and the PLS symbols PLS of the signalframe 41030 or 41040 for rapid service acquisition and data decoding. Ifthe frame structure module configures the PLS offset portion 41031 or41042 in the signal frame, the preamble may include PLS offsetinformation PLS_offset. According to an embodiment of the presentinvention, the value of PLS_offset may be defined as the length of OFDMsymbols used to configure the PLS offset portion.

Due to the PLS offset portion configured in the signal frame, thereceiver may ensure T_Sync corresponding to a time for detecting anddecoding the preamble.

A description is now given of a method for determining the value ofPLS_offset.

The length of an OFDM symbol in the signal frame is defined as T_Symbol.If the signal frame does not include the edge symbol E_(H), the lengthof OFDM symbols including the PLS offset (the value of PLS_offset) maybe determined as a value equal to or greater than a ceiling value (orrounded-up value) of T_Sync/T_Symbol.

If the signal frame includes the edge symbol E_(H), the length of OFDMsymbols including PLS_offset may be determined as a value equal to orgreater than (a ceiling value (or rounded-up value) ofT_Sync/T_Symbol)−1.

Accordingly, the receiver may know of the structure of the receivedsignal frame based on data including the value of PLS_offset which isacquired by detecting and decoding the preamble. If the value ofPLS_offset is 0, it can be noted that the signal frame according to anembodiment of the present invention has a structure in which the PLSsymbols are sequentially mapped subsequently to the preamble symbol.Alternatively, if the value of PLS_offset is 0 and the signal frameincludes the edge symbol, the receiver may know of the signal frame hasa structure in which the edge symbol and the PLS symbols aresequentially mapped subsequently to the preamble symbol.

The frame structure module may configure the PLS offset portion 41031 tobe mapped to the data symbols DATA Frame N or the PLS symbols PLS.Accordingly, as illustrated in FIG. 76(b), the frame structure modulemay allocate data symbols to which data of a previous frame (e.g., FrameN−1) is mapped, to the PLS offset portion. Alternatively, although notshown in FIG. 76(b), the frame structure module may allocate PLS symbolsto which PLS data of a next frame is mapped, to the PLS offset portion.

The frame structure module may perform one or more quantizationoperations on PLS_offset to reduce signaling bits of the preamble.

A description is now given of an example in which the frame structuremodule allocates 2 bits of PLS_offset to the preamble to be signaled.

If the value of PLS_offset is “00”, the length of the PLS offset portionis 0. This means that the PLS data is mapped in the signal frameimmediately next to the preamble or immediately next to the edge symbolif the edge symbol is present.

If the value of PLS_offset is “01”, the length of the PLS offset portionis ¼*L_Frame. Here, L_Frame refers to the number of OFDM symbols whichcan be included in a frame.

If the value of PLS_offset is “10”, the length of the PLS offset portionis 2/4*L_Frame.

If the value of PLS_offset is “11”, the length of the PLS offset portionis ¾*L_Frame.

The above-described method for determining the value of PLS_offset andthe length of the PLS offset portion by the frame structure module ismerely an exemplary embodiment, and terms and values thereof may varyaccording to the intention of a designer.

As described above, FIG. 76 illustrates a frame structure in a case whena time corresponding to a plurality of OFDM symbols (PLS_offset) istaken for synchronization after the preamble is detected and decoded.After the preamble is detected and decoded, the receiver may compensateinteger frequency offset, fractional frequency offset and samplingfrequency offset for a time for receiving a plurality of OFDM symbols(PLS_offset) based on information such as a continual pilot and a guardinterval.

A description is now given of an effect achievable when the framestructure module according to an embodiment of the present inventionensures T_Sync by allocating the PLS offset portion to the signal frame.

If the signal frame includes the PLS offset portion, a reception channelscanning time and a service data acquisition time taken by the receivermay be reduced.

Specifically, PLS information in the same frame as the preamble detectedand decoded by the receiver may be decoded within a time for receivingthe frame, and thus the channel scanning time may be reduced. In futurebroadcast systems, various systems can transmit data in a physical frameusing TDM and thus the complexity of channel scanning is increased. Assuch, if the structure of the signal frame to which the PLS offsetportion is allocated according to an embodiment of the present inventionis used, the channel scanning time may be reduced more.

Further, compared to the structure of the signal frame to which the PLSoffset portion is not allocated (FIG. 76(a)), in the structure of thesignal frame to which the PLS offset portion is allocated (FIG. 76(b)),the receiver may expect a service data acquisition time gaincorresponding to the difference between the length of the signal frameand the length of the PLS_offset portion.

The above-described effect of allocating the PLS offset portion may beachieved in a case when the receiver cannot decode PLS data in the sameframe as the received preamble symbol. If the frame structure module canbe designed to decode the preamble and the edge symbol withoutallocating the PLS offset portion, the value of PLS_offset may be set to0.

FIG. 77 is a flowchart of a broadcast signal transmission methodaccording to another embodiment of the present invention.

A broadcast signal transmission apparatus according to an embodiment ofthe present invention may encode service data for transmitting one ormore broadcast service components (S42000). The broadcast servicecomponents may correspond to broadcast service components for a fixedreceiver and each broadcast service component may be transmitted on aframe basis. The encoding method is as described above.

Then, the broadcast signal transmission apparatus according to anembodiment of the present invention may encode physical signaling datainto an LDPC code based on shortening and puncturing. Here, the physicalsignaling data is encoded based on a code rate determined based on thesize of physical signaling data (S42010). To determine the code rate andencode the physical signaling data by the broadcast signal transmissionapparatus according to an embodiment of the present invention, asdescribed above in relation to FIGS. 36 to 39, the LDPC encoder modulemay LDPC-encode input PLS data or BCH-encoded PLS data based on ashortened/punctured LDPC code and output the LDPC-encoded PLS data. LDPCencoding may be performed based on one of mother code types havingdifferent code rates according to the size of input physical signalingdata including BCH parity.

Then, the broadcast signal transmission apparatus according to anembodiment of the present invention may map the encoded service dataonto constellations (S42020). The mapping method is as described abovein relation to FIGS. 16 to 35.

Then, the broadcast signal transmission apparatus according to anembodiment of the present invention builds at least one signal frameincluding preamble data, the physical signaling data and the mappedservice data (S42030). To build the signal frame by the broadcast signaltransmission apparatus according to an embodiment of the presentinvention, as described above in relation to FIGS. 40 and 41, PLSrepetition for including two or more pieces of physical signaling dataincluding information about two or more frames in a single frame may beused. Further, the broadcast signal transmission apparatus according toan embodiment of the present invention may configure an offset portionin a front part of physical signaling data for a current frame mapped tothe signal frame, and map service data of a previous frame or physicalsignaling data of a next frame to the offset portion.

Then, the broadcast signal transmission apparatus according to anembodiment of the present invention may modulate the built signal frameusing OFDM (S42040).

Then, the broadcast signal transmission apparatus according to anembodiment of the present invention may transmit one or more broadcastsignals carrying the modulated signal frame (S42050).

FIG. 78 is a flowchart of a broadcast signal reception method accordingto another embodiment of the present invention.

The broadcast signal reception method of FIG. 78 corresponds to aninverse procedure of the broadcast signal transmission method describedabove in relation to FIG. 77.

The broadcast signal reception apparatus according to an embodiment ofthe present invention may receive one or more broadcast signals(S43000). Then, the broadcast signal reception apparatus according to anembodiment of the present invention may demodulate the receivedbroadcast signals using OFDM (S43010).

Then, the broadcast signal reception apparatus according to anembodiment of the present invention may parse at least one signal framefrom the demodulated broadcast signals. Here, the signal frame parsedfrom the broadcast signals may include preamble data, physical signalingdata and service data (S43020). To build the signal frame by thebroadcast signal transmission apparatus according to an embodiment ofthe present invention, as described above in relation to FIGS. 75 and76, PLS repetition for including two or more pieces of physicalsignaling data including information about two or more frames in asingle frame may be used. Further, the broadcast signal transmissionapparatus according to an embodiment of the present invention mayconfigure an offset portion in a front part of physical signaling datafor a current frame mapped to the signal frame, and map service data ofa previous frame or physical signaling data of a next frame to theoffset portion. Then, the broadcast signal reception apparatus accordingto an embodiment of the present invention may decode the physicalsignaling data based on LDPC. Here, the physical signaling data is ashortened/punctured LDPC code encoded based on a code rate determinedbased on the size of the physical signaling data (S43030). To determinethe code rate and decode the physical signaling data, as described abovein relation to FIGS. 71 to 74, the LDPC decoder module may LDPC-decodeinput PLS data or BCH-encoded PLS data based on a shortened/puncturedLDPC code and output the LDPC-decoded PLS data. LDPC decoding may beperformed based on different code rates according to the size ofphysical signaling data including BCH parity.

Then, the broadcast signal reception apparatus according to anembodiment of the present invention may demap the service data includedin the signal frame (S43040).

Then, the broadcast signal reception apparatus according to anembodiment of the present invention may decode the service data fortransmitting one or more broadcast service components (S43050).

FIG. 79 illustrates a waveform generation module and a synchronization &demodulation module according to another embodiment of the presentinvention.

FIG. 79(a) shows the waveform generation module according to anotherembodiment of the present invention. The waveform generation module maycorrespond to the aforementioned waveform generation module. The waveform generation module according to another embodiment may include a newreference signal insertion & PAPR reduction block. The new referencesignal insertion & PAPR reduction block may correspond to theaforementioned reference signal insertion & PAPR reduction block.

The present invention provides a method for generating a continuouspilot (CP) pattern inserted into predetermined positions of each signalblock. In addition, the present invention provides a method foroperating CPs using a small-capacity memory (ROM). The new referencesignal insertion & PAPR reduction block according to the presentinvention may operate according to the methods for generating andoperating a CP pattern provided by the present invention.

FIG. 79(b) illustrates a synchronization & demodulation module accordingto another embodiment of the present invention. The synchronization &demodulation module may correspond to the aforementioned synchronization& demodulation module. The synchronization & demodulation module mayinclude a new reference signal detector. The new reference signaldetector may correspond to the aforementioned reference signal detector.

The new reference signal detector according to the present invention mayperform operation of a receiver using CPs according to the method forgenerating and operating CPs, provided by the present invention. CPs maybe used for synchronization of the receiver. The new reference signaldetector may detect a received reference signal to aid insynchronization or channel estimation of the receiver. Here,synchronization may be performed through coarse auto frequency control(AFC), fine AFC and/or common phase error correction (CPE).

At a transmitter, various cells of OFDM symbols may be modulated throughreference information. The reference information may be called a pilot.Pilots may include a SP (scattered pilot), CP (continual pilot), edgepilot, FSS (frame signaling symbol) pilot, FES (frame edge symbol)pilot, etc. Each pilot may be transmitted at a specific boosted powerlevel according to pilot type or pattern.

The CP may be one of the aforementioned pilots. A small quantity of CPsmay be randomly distributed in OFDM symbols and operated. In this case,an index table in which CP position information is stored in a memorymay be efficient. The index table may be referred to as a referenceindex table, a CP set, a CP group, etc. The CP set may be determineddepending on FFT size and SP pattern.

CPs may be inserted into each frame. Specifically, CPs can be insertedinto symbols of each frame. The CPs may be inserted in a CP patternaccording to the index table. However, the size of the index table mayincrease as the SP pattern is diversified and the number of activecarriers (NOC) increases.

To solve this problem, the present invention provides a method foroperating CPs using a small-capacity memory. The present inventionprovides a pattern reversal method and a position multiplexing method.According to these methods, storage capacity necessary for the receivercan be decreased.

The design concept of a CP pattern may be as follows. The number ofactive data carriers (NOA) in each OFDM symbol is held constant. Theconstant NOA may conform to a predetermined NOC (or FFT mode) and SPpattern.

The CP pattern can be changed based on NOC and SP pattern to check thefollowing two conditions: reduction of signaling information; andsimplification of interaction between a time interleaver and carriermapping.

Subsequently, CPs to be positioned in an SP-bearing carrier and anon-SP-bearing carrier can be fairly selected. This selection processmay be carried out for a frequency selective channel. The selectionprocess may be performed such that the CPs are randomly distributed withroughly even distribution over a spectrum. The number of CP positionsmay increase as the NOC increases. This may serve to preserve overheadof the CPs.

The pattern reversal method will now be briefly described. A CP patternthat can be used in an NOC or SP pattern may be generated based on theindex table. CP position values may be arranged into an index tablebased on the smallest NOC. The index table may be referred to as areference index table. Here, the CP position values may be randomlylocated. For a larger NOC, the index table can be extended by reversingthe distribution pattern of the index table. Extension may not beachieved by simple repetition according to a conventional technique.Cyclic shifting may precede reversal of the distribution pattern of theindex table according to an embodiment. According to the patternreversal method, CPs can be operated even with a small-capacity memory.The pattern reversal method may be applied to NOC and SP modes. Inaddition, according to the pattern reversal method, CP positions may beevenly and randomly distributed over the spectrum. The pattern reversalmethod will be described in more detail later.

The position multiplexing method will now be briefly described. Like thepattern reversal method, a CP pattern that can be used in the NOC or SPpattern may be generated based on the index table. First, positionvalues for randomly positioning CPs may be aligned into an index table.This index table may be referred to as a reference index table. Theindex table may be designed in a sufficiently large size to be usedfor/applied to all NOC modes. Then, the index table may be multiplexedthrough various methods such that CP positions are evenly and randomlydistributed over the spectrum for an arbitrary NOC. The positionmultiplexing method will be described in more detail later.

FIG. 80 illustrates definition of a CP bearing SP and a CP not bearingSP according to an embodiment of the present invention.

A description will be given of a random CP position generator prior todescription of the pattern reversal method and the position multiplexingmethod. The pattern reversal method and the position multiplexing methodmay require the random CP position generator.

Several assumptions may be necessary for the random CP positiongenerator. First, it can be assumed that CP positions are randomlyselected by a PN generator at a predetermined NOC. That is, it can beassumed that the CP positions are randomly generated using a PRBSgenerator and provided to the reference index table. It can be assumedthat the NOA in each OFDM symbol is constantly maintained. The NOA ineach OFDM symbol may be constantly maintained by appropriately selectingCP bearing SPs and CP not bearing SPs.

In FIG. 80, uncolored portions represent CP not bearing SPs and coloredportions represent CP bearing SPs.

FIG. 81 shows a reference index table according to an embodiment of thepresent invention.

The reference index table shown in FIG. 81 may be a reference indextable generated using the aforementioned assumptions. The referenceindex table considers 8K FFT mode (NOC: 6817) and SP mode (Dx:2, Dy:4).The index table shown in FIG. 81(a) may be represented as a graph shownin FIG. 81(b).

FIG. 82 illustrates the concept of configuring a reference index tablein CP pattern generation method #1 using the position multiplexingmethod.

A description will be given of CP pattern generation method #1 using theposition multiplexing method.

When a reference index table is generated, the index table can bedivided into sub index tables having a predetermined size. Different PNgenerators (or different seeds) may be used for the sub index tables togenerate CP positions. FIG. 82 shows a reference index table considering8, 16 and 32K FFT modes. That is, in the case of 8K FFT mode, a singlesub index table can be generated by PN1. In the case of 16K FFT mode,two sub index tables can be respectively generated by PN1 and PN2. TheCP positions may be generated based on the aforementioned assumptions.

For example, when the 16K FFT mode is supported, CP position valuesobtained through a PN1 and PN2 generator can be sequentially arranged todistribute all CP positions. When the 32K FFT mode is supported, CPposition values obtained through a PN3 and PN4 generator can beadditionally arranged to distribute all CP positions.

Accordingly, CPs can be evenly and randomly distributed over thespectrum. In addition, a correlation property between CP positions canbe provided.

FIG. 83 illustrates a method for generating a reference index table inCP pattern generation method #1 using the position multiplexing methodaccording to an embodiment of the present invention.

In the present embodiment, CP position information may be generated inconsideration of an SP pattern with Dx=3 and Dy=4. In addition, thepresent embodiment may be implemented in 8K/16K/32K FFT modes (NOC:1817/13633/27265).

CP position values may be stored in a sub index table using the 8K FFTmode as a basic mode. When 16K or higher FFT modes are supported, subindex tables may be added to the stored basic sub index table. Values ofthe added sub index tables may be obtained by adding a predeterminedvalue to the stored basic sub index table or shifting the basic subindex table.

CP position values provided to the ends of sub index tables PN1, PN2 andPN3 may refer to values necessary when the corresponding sub indextables are extended. That is, the CP position values may be values formultiplexing. The CP position values provided to the ends of the subindex tables are indicated by ovals in FIG. 83.

The CP position values v provided to the ends of the sub index tablesmay be represented as follows.

v=i·D _(x) ·D _(y)  [Math FIG. 11]

Here, v can be represented as an integer multiple i of D_(x)·D_(y). Whenthe 8K FFT mode is applied, the last position value of sub index tablePN1 may not be applied. When the 16K FFT mode is applied, the lastposition value of sub index table PN1 is applied whereas the lastposition value of sub index table PN2 may not be applied. Similarly,when the 32K FFT mode is applied, all the last position values of subindex tables PN1, PN2 and PN3 may be applied.

In CP pattern generation method #1 using the position multiplexingmethod, the aforementioned multiplexing rule can be represented by thefollowing equation. The following equation may be an equation forgenerating CP positions to be used in each FFT mode from a predeterminedreference index table.

                                   [Math  Figure  12]  CP_8K(k) = PN 1(k),   for  1 ≤ k ≤ S_(PN 1) − 1${{CP\_}16{K(k)}} = \left\{ {{\begin{matrix}{{{PN}\; 1(k)},} & {{{if}\mspace{14mu} 1} \leq k \leq S_{{PN}\; 1}} \\{{\alpha_{1} + {{PN}\; 2\left( {k - S_{{PN}\; 1}} \right)}},} & {{{{elseif}\mspace{14mu} S_{{PN}\; 1}} + 1} \leq k \leq {S_{{PN}\; 12} - 1}}\end{matrix}{CP\_}32{K(k)}} = \left\{ {{\begin{matrix}{{{PN}\; 1(k)},} & {{{if}\mspace{14mu} 1} \leq k \leq S_{{PN}\; 1}} \\{{\alpha_{1} + {{PN}\; 2\left( {k - S_{{PN}\; 1}} \right)}},} & {{{{elseif}\mspace{14mu} S_{{PN}\; 1}} + 1} \leq k \leq S_{{PN}\; 12}} \\{{\alpha_{2} + {{PN}\; 3\left( {k - S_{{PN}\; 12}} \right)}},} & {{{{elseif}\mspace{14mu} S_{{PN}\; 12}} + 1} \leq k \leq S_{{PN}\; 123}} \\{{\alpha_{3} + {{PN}\; 4\left( {k - S_{{PN}\; 123}} \right)}},} & {{{{elseif}\mspace{14mu} S_{{PN}\; 123}} + 1} \leq k \leq S_{{PN}\; 1234}}\end{matrix}\mspace{20mu} {where}\mspace{14mu} S_{{PN}\; 12}} = {{S_{{PN}\; 1} + {S_{{PN}\; 2}\mspace{20mu} S_{{PN}\; 123}}} = {{S_{{PN}\; 1} + S_{{PN}\; 2} + {S_{{PN}\; 3}\mspace{20mu} S_{{PN}\; 1234}}} = {S_{{PN}\; 1} + S_{{PN}\; 2} + S_{{PN}\; 3} + S_{{PN}\; 4}}}}} \right.} \right.$

Math FIG. 12 may be an equation for generating CP position values to beused in each FFT mode based on the predetermined reference index table.Here, CP_8/16/32K respectively denote CP patterns in 8K, 16K and 32K FFTmodes and PN_1/2/3/4 denote sub index table names. S_(PN) _(_)_(1/2/3/4) respectively represent the sizes of sub index tables PN1,PN2, PN3 and PN4 and α_(1/2/3) represent shifting values for evenlydistributing added CP positions.

In CP_8K(k) and CP_16K(k), k is limited to S_(PN1)−1 and S_(PN12)−1.Here, −1 is added since the last CP position value v is excluded, asdescribed above.

FIG. 84 illustrates the concept of configuring a reference index tablein CP pattern generation method #2 using the position multiplexingmethod according to an embodiment of the present invention.

CP pattern generation method #2 using the position multiplexing methodwill now be described.

CP pattern generation method #2 using the position multiplexing methodmay be performed in a manner that a CP pattern according to FFT mode issupported. CP pattern generation method #2 may be performed in such amanner that PN1, PN2, PN3 and PN4 are multiplexed to support a CP suitedto each FFT mode. Here, PN1, PN2, PN3 and PN4 are sub index tables andmay be composed of CP positions generated by different PN generators.PN1, PN2, PN3 and PN4 may be assumed to be sequences in which CPposition values are distributed randomly and evenly. While the referenceindex table may be generated through a method similar to theaforementioned CP pattern generation method #1 using the positionmultiplexing method, a detailed multiplexing method may differ from CPpattern generation method #1.

A pilot density block can be represented as N_(blk). The number ofallocated pilot density blocks N_(blk) may depend on FFT mode in thesame bandwidth. That is, one pilot density block N_(blk) may beallocated in the case of 8K FFT mode, two pilot density blocks N_(blk)may be allocated in the case of 16K FFT mode and four pilot densityblocks N_(blk) may be allocated in the case of 32K FFT mode. PN1 to PN4may be multiplexed in an allocated region according to FFT mode togenerate CP patterns.

PN1 to PN4 may be generated such that a random and even CP distributionis obtained. Accordingly, the influence of an arbitrary specific channelmay be mitigated. Particularly, PN1 can be designed such thatcorresponding CP position values are disposed in the same positions inphysical spectrums of 8K, 16K and 32K. In this case, a receptionalgorithm for synchronization can be implemented using simple PN1.

In addition, PN1 to PN4 may be designed such that they have excellentcross correlation characteristics and auto correlation characteristics.

In the case of PN2 in which CP positions are additionally determined inthe 16K FFT mode, the CP positions can be determined such that PN2 hasexcellent auto correlation characteristics and even distributioncharacteristics with respect to the position of PN1 determined in the 8KFFT mode. Similarly, in the case of PN3 and PN4 in which CP positionsare additionally determined in the 32K FFT mode, the CP positions can bedetermined such that auto correlation characteristics and evendistribution characteristics are optimized based on the positions of PN1and PN2 determined in 16K FFT mode.

CPs may not be disposed in predetermined portions of both edges of thespectrum. Accordingly, it is possible to mitigate loss of some CPs whenan integral frequency offset (ICFO) is generated.

FIG. 85 illustrates a method for generating a reference index table inCP pattern generation method #2 using the position multiplexing method.

PN1 can be generated in case of the 8K FFT mode, PN1 and PN2 can begenerated in case of the 16K FFT mode and PN1, PN2, PN3 and PN4 can begenerated in case of the 32K FFT mode. The generation process may beperformed according to a predetermined multiplexing rule.

FIG. 85 illustrates that two pilot density blocks N_(blk) in case of the16K FFT mode and four pilot density blocks N_(blk) in case of the 32KFFT mode can be included in a region which can be represented by asingle pilot density block N_(blk)* on the basis of the 8K FFT mode. PNsgenerated according to each FFT mode can be multiplexed to generate a CPpattern.

In the case of 8K FFT mode, a CP pattern can be generated using PN1.That is, PN1 may be a CP pattern in the 8K FFT mode.

In the case of 16K FFT mode, PN1 can be positioned in the first pilotdensity block (first N_(blk)) and PN2 can be disposed in the secondpilot density block (second N_(blk)) to generate a CP pattern.

In the case of 32K FFT mode, PN1 can be disposed in the first pilotdensity block (first N_(blk)), PN2 can be disposed in the second pilotdensity block (second N_(blk)), PN3 can be disposed in the third pilotdensity block (third N_(blk)) and PN4 can be disposed in the fourthpilot density block (fourth N_(blk)) to generate a CP pattern. WhilePN1, PN2, PN3 and PN4 are sequentially disposed in the presentembodiment, PN2 may be disposed in the third pilot density block (thirdN_(blk)) in order to insert CPs into similar positions of the spectrumas in the 16K FFT mode.

In CP pattern generation method #2 using the position multiplexingmethod, the aforementioned multiplexing rule can be represented by thefollowing equation. The following equation may be an equation forgenerating CP positions to be used in each FFT mode from a predeterminedreference index table.

$\begin{matrix}{\mspace{79mu} {{{{CP\_}8K} = {{PN}\; 1(k)}},{{{CP\_}16{K(k)}} = \left\{ {{\begin{matrix}{{{PN}\; 1\left( {{{{ceil}\left( \frac{k}{2N_{blk}} \right)} \cdot N_{blk}} + {{mod}\left( {k,{2N_{blk}}} \right)}} \right)},} & {0 \leq {{mod}\left( {k,{2N_{blk}}} \right)} < N_{blk}} \\{{{PN}\; 2\left( {{{{ceil}\left( \frac{k}{2N_{blk}} \right)} \cdot N_{blk}} + {{mod}\left( {\left( {k - N_{blk}} \right),{2N_{blk}}} \right)}} \right)},} & {N_{blk} \leq {{mod}\left( {k,{2N_{blk}}} \right)} < {2N_{blk}}}\end{matrix}{CP\_}32{K(k)}} = \left\{ \begin{matrix}{{{PN}\; 1\left( {{{{ceil}\left( \frac{k}{4N_{blk}} \right)} \cdot N_{blk}} + {{mod}\left( {k,{4N_{blk}}} \right)}} \right)},} & {0 \leq {{mod}\left( {k,{4N_{blk}}} \right)} < N_{blk}} \\{{{PN}\; 2\left( {{{{ceil}\left( \frac{k}{4N_{blk}} \right)} \cdot N_{blk}} + {{mod}\left( {\left( {k - N_{blk}} \right),{4N_{blk}}} \right)}} \right)},} & {N_{blk} \leq {{mod}\left( {k,{4N_{blk}}} \right)} < {2N_{blk}}} \\{{{PN}\; 3\left( {{{{ceil}\left( \frac{k}{4N_{blk}} \right)} \cdot N_{blk}} + {{mod}\left( {\left( {k - {2N_{blk}}} \right),{4N_{blk}}} \right)}} \right)},} & {{2N_{blk}} \leq {{mod}\left( {k,{4N_{blk}}} \right)} < {3N_{blk}}} \\{{{PN}\; 4\left( {{{{ceil}\left( \frac{k}{4N_{blk}} \right)} \cdot N_{blk}} + {{mod}\left( {\left( {k - {3N_{blk}}} \right),{4N_{blk}}} \right)}} \right)},} & {{3N_{blk}} \leq {{mod}\left( {k,{4N_{blk}}} \right)} < {4N_{blk}}}\end{matrix} \right.} \right.}}} & \left\lbrack {{Math}\mspace{14mu} {Figure}\mspace{14mu} 13} \right\rbrack\end{matrix}$

Math FIG. 13 may be an equation for generating CP position values to beused in each FFT mode based on the predetermined reference index table.Here, CP_8/16/32K respectively denote CP patterns in 8K, 16K and 32K FFTmodes and PN1 to PN4 denote sequences. These sequences may be fourpseudo random sequences. In addition, ceil(X), ceiling function of X,represents a function outputting a minimum value from among integersequal to or greater than X and mod(X,N) is a modulo function capable ofoutputting a remainder obtained when X is divided by N.

For the 16K FFT mode and the 32K FFT mode, sequences PN1 to PN4 may bemultiplexed in offset positions determined according to each FFT mode.In the above equation, offset values may be represented by modulooperation values of predetermined integer multiples of basic N_(blk).The offset values may be different values.

FIG. 86 illustrates a method for generating a reference index table inCP pattern generation method #3 using the position multiplexing methodaccording to an embodiment of the present invention.

In the present embodiment, PN1 to PN4 may be assumed to be sequences inwhich CP position values are distributed randomly and evenly. Inaddition, PN1 to PN4 may be optimized to satisfy correlation and evendistribution characteristics for 8K, 16K and 32K, as described above.

The present embodiment may relate to a scattered pilot pattern forchannel estimation. In addition, the present embodiment may relate to acase in which distance Dx in the frequency direction is 8 and distanceDy in the time direction is 2. The present embodiment may be applicableto other patterns.

As described above, PN1 can be generated in the case of 8K FFT mode, PN1and PN2 can be generated in the case of 16K FFT mode and PN1, PN2, PN3and PN4 can be generated in the case of 32K FFT mode. The generationprocess may be performed according to a predetermined multiplexing rule.

FIG. 86 shows that two pilot density blocks N_(blk) in case of the 16KFFT mode and four pilot density blocks N_(blk) in case of the 32K FFTmode can be included in a region which can be represented by a singlepilot density block N_(blk) on the basis of the 8K FFT mode.

PNs generated according to each FFT mode can be multiplexed to generatea CP pattern. In each FFT mode, CPs may be disposed overlapping with SPs(SP bearing) or disposed not overlapping with SPs (non-SP bearing). Inthe present embodiment, a multiplexing rule for SP bearing or non-SPbearing CP positioning can be applied in order to dispose pilots in thesame positions in the frequency domain.

In the case of SP bearing, PN1 to PN4 may be disposed such that CPpositions are distributed randomly and evenly for an SP offset pattern.Here, PN1 to PN4 may be sequences forming an SP bearing set. PN1 to PN4may be positioned according to the multiplexing rule for each FFT mode.That is, in the case of 16K FFT mode, PN2 added to PN1 can be disposedin positions other than an SP offset pattern in which PN1 is positioned.A position offset with respect to PN2 may be set such that PN2 ispositioned in positions other than the SP offset pattern in which PN1 ispositioned or PN2 may be disposed in a pattern determined through arelational expression. Similarly, in the case of 32K FFT mode, PN3 andPN4 may be configured to be disposed in positions other than SP offsetpatterns in which PN1 and PN2 are positioned.

In case of non-SP bearing, PN1 to PN4 may be positioned according to arelational expression. Here, PN1 to PN4 may be sequences forming anon-SP bearing set.

In CP pattern generation method #3 using the position multiplexingmethod, the aforementioned multiplexing rule can be represented by thefollowing equations. The following equations may be equations forgenerating CP positions to be used in each FFT mode from a predeterminedreference index table.

$\begin{matrix}{{\left. \mspace{79mu} 1 \right)\mspace{14mu} {SP}\mspace{14mu} {bearing}\mspace{14mu} {set}\text{:}}\mspace{20mu} {{{PN}\; 1_{sp}(k)},\mspace{20mu} {{PN}\; 2_{sp}(k)},\mspace{20mu} {{PN}\; 3_{sp}(k)},\mspace{20mu} {{PN}\; 4_{sp}(k)}}\mspace{20mu} {{{{CP}_{sp}\_ 8{K(k)}} = {{PN}\; 1_{sp}(k)}},\mspace{20mu} {{{CP}_{sp}\_ 16{K(k)}} = \left\{ {{\begin{matrix}{{{PN}\; 1_{sp}(k) \times 2},} \\{{{{PN}\; 2_{sp}(k) \times 2} + \alpha_{16\; K}},}\end{matrix}{CP}_{sp}\_ 32{K(k)}} = \left\{ \begin{matrix}{{{CP\_}16{K(k)}*2} = \left\{ \begin{matrix}{\left( {{PN}\; 1_{sp}(k) \times 2} \right) \times 2} \\{\left( {{{PN}\; 1_{sp}(k) \times 2} + \alpha_{16K}} \right) \times 2}\end{matrix} \right.} \\{{{PN}\; 3_{sp}(k)*4} + {\alpha \; 1_{32K}}} \\{{{PN}\; 4_{sp}(k)*4} + {\alpha \; 2_{32K}}}\end{matrix} \right.} \right.}}} & \left\lbrack {{Math}\mspace{14mu} {Figure}\mspace{14mu} 14} \right\rbrack \\{{\left. \mspace{79mu} 2 \right)\mspace{14mu} {Non}\mspace{14mu} {SP}\mspace{14mu} {bearing}\mspace{14mu} {set}\text{:}}\; \mspace{20mu} {{{PN}\; 1_{nonsp}(k)},\mspace{20mu} {{PN}\; 2_{nonsp}(k)},\mspace{20mu} {{PN}\; 3_{nonsp}(k)},\mspace{20mu} {{PN}\; 4_{nonsp}(k)}}\mspace{20mu} {{{{CP}_{nonsp}\_ 8{K(k)}} = {{PN}\; 1_{nonsp}(k)}},\mspace{20mu} {{{CP}_{nonsp}\_ 16{K(k)}} = \left\{ {{\begin{matrix}{{{PN}\; 1_{nonsp}(k) \times 2},} \\{{{{PN}\; 2_{nonsp}(k) \times 2} + \beta_{16K}},}\end{matrix}{CP}_{nonsp}\_ 32{K(k)}} = \left\{ \begin{matrix}{{{CP}_{nonsp}\_ 16{K(k)}*2} = \left\{ \begin{matrix}{\left( {{PN}\; 1_{nonsp}(k) \times 2} \right) \times 2} \\{\left( {{{PN}\; 1_{nonsp}(k) \times 2} + \beta_{16K}} \right) \times 2}\end{matrix} \right.} \\{{{PN}\; 3_{nonsp}(k)*4} + {\beta \; 1_{32K}}} \\{{{PN}\; 4_{nonsp}(k)*4} + {\beta 2}_{32K}}\end{matrix} \right.} \right.}}} & \left\lbrack {{Math}\mspace{14mu} {Figure}\mspace{14mu} 15} \right\rbrack \\{\mspace{79mu} {{{{CP\_}8{K(k)}} = \left\{ {{{CP}_{sp}\_ 8{K(k)}},{{CP}_{nonsp}\_ 8{K(k)}}} \right\}}\mspace{20mu} {{{CP\_}16{K(k)}} = \left\{ {{{CP}_{sp}\_ 16{K(k)}},{{CP}_{nonsp}\_ 16{K(k)}}} \right\}}\mspace{20mu} {{{CP\_}32{K(k)}} = \left\{ {{{CP}_{sp}\_ 32{K(k)}},{{CP}_{nonsp}\_ 32{K(k)}}} \right\}}}} & \left\lbrack {{Math}\mspace{14mu} {Figure}\mspace{14mu} 16} \right\rbrack\end{matrix}$

The above equations may be equations for generating CP position valuesto be used in each FFT mode based on the predetermined reference indextable. Here, CP_8/16/32K respectively denote CP patterns in 8K, 16K and32K FFT modes and CP_(sp) _(_)8/16/32K respectively denote SP bearing CPpatterns in 8K, 16K and 32K FFT modes. CP_(nonsp) _(_)CP 8/16/32Krespectively represent non-SP bearing CP patterns in 8K, 16K and 32K FFTmodes and PN1_(sp), PN2_(sp), PN3_(sp) and PN4_(sp) represent sequencesfor SP bearing pilots. These sequences may be four pseudo randomsequences. These sequences may be included in an SP being set.PN1_(nonsp), PN2_(nonsp), PN3_(nonsp) and PN4_(nonsp) denote sequencesfor non-SP bearing pilots. These sequences may be four pseudo randomsequences and may be included in a non-SP bearing set. In addition,α_(16K), α1_(32K), α2_(32K), β_(16K), β1_(32K) and β2 _(32K) representCP position offsets.

Respective SP bearing CP patterns can be generated using PN1_(sp),PN2_(sp), PN3_(sp)and PN4_(sp), as represented by Math FIG. 14.Respective non-SP bearing patterns can be generated using PN1_(nonsp),PN2_(nonsp), PN3_(nonsp) and PN4_(nonsp), as represented by Math FIG.15. As represented by Math FIG. 16, the CP pattern of each FFT mode canbe composed of an SP bearing CP pattern and a non-SP bearing CP pattern.That is, an SP bearing CP index table can be added to a non-SP bearingCP index table to generate a reference index table. Consequently, CPinsertion can be performed according to the non-SP bearing CP indextable and the SP bearing CP index table. Here, non-SP bearing CPposition values may be called a common CP set and SP bearing CP positionvalues may be called an additional CP set.

CP position offsets may be values predetermined for multiplexing, asdescribed above. The CP position offsets may be allocated to the samefrequency irrespective of FFT mode or used to correct CPcharacteristics.

FIG. 87 illustrates the concept of configuring a reference index tablein CP pattern generation method #1 using the pattern reversal method.

CP pattern generation method #1 using the pattern reversal method willnow be described.

As described above, when the reference index table is generated, thetable can be divided into sub index tables having a predetermined size.The sub index tables may include CP positions generated using differentPN generators (or different seeds).

In the pattern reversal method, two sub index tables necessary in the8K, 16K and 32K FFT modes can be generated by two different PNgenerators. Two sub index tables additionally necessary in the 32K FFTmode can be generated by reversing the pre-generated two sub indextables.

That is, when the 16K FFT mode is supported, CP positions according toPN1 and PN2 can be sequentially arranged to obtain a CP positiondistribution. When the 32K FFT mode is supported, however, CP positionsaccording to PN1 and PN2 can be reversed to obtain a CP positiondistribution.

Accordingly, a CP index table in the 32K FFT mode can include a CP indextable in the 16K FFT mode. In addition, the CP index table in the 16KFFT mode can include a CP index table in the 8K FFT mode. According toan embodiment, the CP index table in the 32K FFT mode may be stored andthe CP index tables in the 8K and 16K FFT modes may beselected/extracted from the CP index table in the 32K FFT mode togenerate the CP index tables in the 8K and 16K FFT modes.

According to the aforementioned pattern reversal method, CP positionscan be distributed evenly and randomly over the spectrum. In addition,the size of a necessary reference index table can be reduced compared tothe aforementioned position multiplexing method. Furthermore, memorystorage capacity necessary for the receiver can be decreased.

FIG. 88 illustrates a method for generating a reference index table inCP pattern generation method #1 using the pattern reversal methodaccording to an embodiment of the present invention.

In the present embodiment, CP position information may be generated inconsideration of an SP pattern with Dx=3 and Dy=4. In addition, thepresent embodiment may be implemented in 8K/16K/32K FFT modes (NOC:1817/13633/27265).

CP position values may be stored in a sub index table using the 8K FFTmode as a basic mode. When 16K or higher FFT modes are supported, subindex tables may be added to the stored basic sub index table. Values ofthe added sub index tables may be obtained by adding a predeterminedvalue to the stored basic sub index table or shifting the basic subindex table.

The 32K FFT mode index table can be generated using sub index tablesobtained by reversing sub index tables of PN1 and PN2.

CP position values provided to the ends of sub index tables PN1 and PN2may refer to values necessary when the corresponding sub index tablesare extended. That is, the CP position values may be values formultiplexing. The CP position values provided to the ends of the subindex tables are indicated by ovals in FIG. 83.

The CP position values v provided to the ends of the sub index tablesmay be represented as follows.

v=i·D _(x) ·D _(y)  [Math FIG. 17]

Here, v can be represented as an integer multiple i of D_(x)·D_(y). Whenthe 8K FFT mode is applied, the last position value of sub index tablePN1 may not be applied. When the 16K FFT mode is applied, the lastposition value of sub index table PN1 is applied whereas the lastposition value of sub index table PN2 may not be applied.

The index table for the 32K FFT mode can be generated using the indextable for the 16K FFT mode and an index table obtained by reversing theindex table for the 16K FFT mode. Accordingly, the last position valueof sub index table PN1 can be used twice and the last position value ofsub index table PN2 can be used only once.

In the extension of a sub index table, extension according to v may benecessary or unnecessary according to embodiment. That is, there may bean embodiment of extending/reversing a sub index table without v.

In CP pattern generation method #1 using the pattern reversal method,the aforementioned multiplexing rule can be represented by the followingequation. The following equation may be an equation for generating CPpositions to be used in each FFT mode from a predetermined referenceindex table.

$\begin{matrix}{\mspace{79mu} {{{{{CP\_}8{K(k)}} = {{PN}\; 1(k)}},\mspace{20mu} {for}}\mspace{20mu} {1 \leq k \leq {S_{{PN}\; 1} - 1}}{{{CP\_}16{K(k)}} = \left\{ {{\begin{matrix}{{{PN}\; 1(k)},} & {{{if}\mspace{14mu} 1} \leq k \leq S_{{PN}\; 1}} \\{{\alpha_{1} + {{PN}\; 2\left( {k - S_{{PN}\; 1}} \right)}},} & {{{{elseif}\mspace{14mu} S_{{PN}\; 1}} + 1} \leq k \leq {S_{{PN}\; 12} - 1}}\end{matrix}{CP\_}32{K(k)}} = \left\{ {{\begin{matrix}{{{PN}\; 1(k)},} & {{{if}\mspace{14mu} 1} \leq k \leq S_{{PN}\; 1}} \\{{\alpha_{1} + {{PN}\; 2\left( {k - S_{{PN}\; 1}} \right)}},} & {{{{elseif}\mspace{14mu} S_{{PN}\; 1}} + 1} \leq k \leq {S_{{PN}\; 12} - 1}} \\{{\alpha_{2} + \left( {\beta - {{PN}\; 1\left( {k - S_{{PN}\; 12} + 1} \right)}} \right)},} & {{{elseif}\mspace{14mu} S_{{PN}\; 12}} \leq k \leq {S_{{PN}\; 121} - 1}} \\{{\alpha_{3} + \left( {\beta - {{PN}\; 2\left( {k - S_{{PN}\; 121} + 1} \right)}} \right)},} & {{{elseif}\mspace{14mu} S_{{PN}\; 121}} \leq k \leq {S_{{PN}\; 1212} - 1}}\end{matrix}\mspace{20mu} {where}\mspace{20mu} S_{{PN}\; 12}} = {{S_{{PN}\; 1} + {S_{{PN}\; 2}\mspace{20mu} S_{{PN}\; 121}}} = {{{2S_{{PN}\; 1}} + {S_{{PN}\; 2}\mspace{20mu} S_{{PN}\; 1212}}} = {{{2S_{{PN}\; 1}} + {2S_{{PN}\; 2}\mspace{20mu} \beta}} = {{aD}_{x}D_{y}}}}}} \right.} \right.}}} & \left\lbrack {{Math}\mspace{14mu} {Figure}\mspace{14mu} 18} \right\rbrack\end{matrix}$

A CP pattern in each FFT mode can be generated according to Math FIG.18. Here, symbols may be the same as the above-described ones. β denotesan integer closest to the NOA of the 8K FFT mode. That is, when the NOAis 6817, β may be 6816.

In CP_8K(k), CP_16K(k) and CP_32K(k), k may be respectively limited toS_(PN1)−1, S_(PN12)−1, S_(PN121)−1 and S_(PN1212)−1. Here, −1 is addedsince the last CP position value v may be excluded according tosituation, as described above. In Math FIG. 18,

β−PN1(k−S _(PN12)+1)).

β−PN2(k−S _(PN121)+1)),

in a box represents pattern reversal.

FIG. 89 illustrates the concept of configuring a reference index tablein CP pattern generation method #2 using the pattern reversal methodaccording to an embodiment of the present invention.

CP pattern generation method #2 using the pattern reversal method willnow be described.

As described above, when the reference index table is generated, thetable can be divided into sub index tables having a predetermined size.The sub index tables may include CP positions generated using differentPN generators (or different seeds).

Two sub index tables necessary in the 8K, 16K and 32K FFT modes can begenerated by two different PN generators, as described above. Two subindex tables additionally necessary in the 32K FFT mode can be generatedby reversing the pre-generated two sub index tables. However, CP patterngeneration method #2 using the pattern reversal method can generate twonecessary sub index tables by cyclic-shifting patterns and thenreversing the patterns rather than simply reversing the previouslygenerated two sub index tables. Reversing operation may precede cyclicshifting operation according to embodiment. Otherwise, simple shiftinginstead of cyclic shifting may be performed according to embodiment.

Accordingly, a CP index table in the 32K FFT mode can include a CP indextable in the 16K FFT mode. In addition, the CP index table in the 16KFFT mode can include a CP index table in the 8K FFT mode. According toan embodiment, the CP index table in the 32K FFT mode may be stored andthe CP index tables in the 8K and 16K FFT modes may beselected/extracted from the CP index table in the 32K FFT mode togenerate the CP index tables in the 8K and 16K FFT modes.

As described above, when the 16K FFT mode is supported, CP positionvalues according to PN1 and PN2 can be sequentially arranged to obtain aCP position distribution. However, according to CP pattern generationmethod #2 using the pattern reversal method, CP position valuesaccording to PN1 and PN2 can be cyclically shifted and then reversed toobtain a CP position distribution when the 32K FFT mode is supported.

According to CP pattern generation method #2 using the pattern reversalmethod, CP positions can be distributed evenly and randomly over thespectrum. In addition, the size of a necessary reference index table canbe reduced compared to the aforementioned position multiplexing method.Furthermore, memory storage capacity necessary for the receiver can bedecreased.

In CP pattern generation method #2 using the pattern reversal method,the aforementioned multiplexing rule can be represented by the followingequation. The following equation may be an equation for generating CPpositions to be used in each FFT mode from a predetermined referenceindex table.

$\begin{matrix}{\mspace{79mu} {{{{{CP\_}8{K(k)}} = {{PN}\; 1(k)}},\mspace{20mu} {for}}\mspace{20mu} {1 \leq k \leq {S_{{PN}\; 1} - 1}}{{{CP\_}16{K(k)}} = \left\{ {{\begin{matrix}{{{PN}\; 1(k)},} & {{{if}\mspace{14mu} 1} \leq k \leq S_{{PN}\; 1}} \\{{\alpha_{1} + {{PN}\; 2\left( {k - S_{{PN}\; 1}} \right)}},} & {{{{elseif}\mspace{14mu} S_{{PN}\; 1}} + 1} \leq k \leq {S_{{PN}\; 12} - 1}}\end{matrix}{CP\_}32{K(k)}} = \left\{ {{\begin{matrix}{{{PN}\; 1(k)},} & {{{if}\mspace{14mu} 1} \leq k \leq S_{{PN}\; 1}} \\{{\alpha_{1} + {{PN}\; 2\left( {k - S_{{PN}\; 1}} \right)}},} & {{{{elseif}\mspace{14mu} S_{{PN}\; 1}} + 1} \leq k \leq {S_{{PN}\; 12} - 1}} \\{{{mod}\left( {{\gamma_{1} + \alpha_{2} + \left( {\beta - {{PN}\; 1\left( {k - S_{{PN}\; 12} + 1} \right)}} \right)},\beta} \right)},} & {{{elseif}\mspace{14mu} S_{{PN}\; 12}} \leq k \leq {S_{{PN}\; 121} - 1}} \\{{{mod}\left( {{\gamma_{2} + \alpha_{3} + \left( {\beta - {{PN}\; 2\left( {k - S_{{PN}\; 121} + 1} \right)}} \right)},\beta} \right)},} & {{{elseif}\mspace{14mu} S_{{PN}\; 121}} \leq k \leq {S_{{PN}\; 1212} - 1}}\end{matrix}\mspace{20mu} {where}\mspace{20mu} S_{{PN}\; 12}} = {{S_{{PN}\; 1} + {S_{{PN}\; 2}\mspace{20mu} S_{{PN}\; 121}}} = {{{2S_{{PN}\; 1}} + {S_{{PN}\; 2}\mspace{20mu} S_{{PN}\; 1212}}} = {{{2S_{{PN}\; 1}} + {2S_{{PN}\; 2}\mspace{20mu} \beta}} = {{aD}_{x}D_{y}}}}}} \right.} \right.}}} & \left\lbrack {{Math}\mspace{14mu} {Figure}\mspace{14mu} 19} \right\rbrack\end{matrix}$

A CP pattern in each FFT mode can be generated according to Math FIG.19. Here, symbols may be the same as the above-described ones. β denotesan integer closest to the NOA of the 8K FFT mode. That is, when the NOAis 6817, β may be 6816. γ_(1/2) is a cyclic shift value.

In CP_8K(k), CP_16K(k) and CP_32K(k), k may be respectively limited toS_(PN1)−1, S_(PN12)−1, S_(PN121)−1 and S_(PN1212)−1. Here, −1 is addedsince the last CP position value v may be excluded according tosituation, as described above. In Math FIG. 19,

mod(γ₁+α₂+(β−PN1(k−S _(PN12)+1)),β),

mod(γ₂+α₁+(β−PN2(k−S _(PN121)+1)),β),

in a box represents pattern reversal and cyclic shifting.

The CP pattern can be generated by a method other than aforementioned CPpattern generation methods. According to other embodiments, a CP set (CPpattern) of certain FFT size can be generated from a CP set of other FFTsize, organically and dependently. In this case, a whole CP set or apart of the CP set can be base of generation process. For example, a CPset of 16K FFT mode can be generated by selecting/extracting CPpositions from a CP set of 32K FFT mode. In same manner, a CP set of 8KFFT mode can be generated by selecting/extracting CP positions from a CPset of 32K FFT mode.

According to other embodiments, CP set can include SP bearing CPpositions and/or non SP bearing CP positions. Non SP bearing CPpositions can be referred to as common CP set. SP bearing CP positionscan be referred to as additional CP set. That is, CP set can include acommon CP set and/or an additional CP set. A case that only a common CPset is included in the CP set can be referred to as normal CP mode. Acase that the CP set includes both a common CP set and an additional CPset can be referred to as extended CP mode.

Values of common CP sets can be different based on FFT size. Accordingto embodiments, the common CP set can be generated by aforementionedPattern reversal method and/or Position multiplexing method.

Values of additional CP sets can be different based on transmissionmethods, such as SISO or MIMO. In situation that additional robustnessis needed, such as mobile reception, or for any other reasons,additional CP positions can be added to the CP set, by adding anadditional CP set.

Consequently, CP insertion can be performed according to the CP set(reference index table).

As described above, the broadcast signal transmission apparatusaccording to an embodiment or the above-mentioned waveform transformblock 7200 may insert pilots into a signal frame generated from a framestructure module 1200, and may OFDM-modulate broadcast signals usingtransmission (Tx) parameters. Tx parameters according to the embodimentmay also be called OFDM parameters.

The present invention proposes Tx parameters that can satisfy a spectrummask reference contained in a transmission (Tx) band for the nextgeneration broadcast transmission/reception (Tx/Rx) system, can maximizeTx efficiency, and can be applied to a variety of Rx scenarios.

FIG. 90 shows a table illustrating information related to a receptionmode according to an embodiment of the present invention.

A Table shown in FIG. 90 may include a network configuration accordingto a reception mode of the next generation broadcast Tx/Rx system.

As described above, the reception modes according to the embodiment canbe classified into a Fixed Rooftop environment and a Handheld portableenvironment, and a representative channel for each environment can bedecided.

In addition, the broadcast signal transmission apparatus according tothe embodiment can decide the transmission (Tx) mode according to theabove-mentioned reception mode. That is, the broadcast signaltransmission apparatus according to the embodiment may process broadcastservice data using the non-MIMO schemes (MISO and SISO schemes) or theMIMO scheme according to the broadcast service characteristics (i.e.,according to the reception mode). Accordingly, the broadcast signal foreach Tx mode may be transmitted and received through a Tx channelcorresponding to the corresponding processing scheme.

In this case, according to one embodiment of the present invention,broadcast signals of individual Tx modes can be identified andtransmitted in units of a signal frame. In addition, each signal framemay include a plurality of OFDM symbols. Each OFDM symbol may becomprised of the above-mentioned preamble (or preamble symbols) and aplurality of data symbols configured to transmit data corresponding to abroadcast signal.

A left column of the Table shown in FIG. 90 shows the above-mentionedthree reception modes.

In case of the fixed rooftop environment, the broadcast signal receptionapparatus may receive broadcast signals through the rooftop antennalocated at the height of 10 ms or higher above the ground. Accordingly,since a direct path can be guaranteed, a Rician channel isrepresentatively used, the Rician channel is less affected by Doppler,and the range of a delay spread may be limited according to the use of adirectional antenna.

In case of the handheld portable environment and the handheld mobileenvironment, the broadcast signal reception apparatus may receivebroadcast signals through the omi-directional antenna located at theheight of 1.5 m or less above the ground. In this case, a Rayleighchannel may be representatively used as the Tx channel environment basedon reflected waves, and may obtain the range of a delay spread of achannel longer than the directional antenna.

In case of the handheld portable environment, a low-level Dopplerenvironment can be supported as the indoor/outdoor receptionenvironments in consideration of mobility such as an adult walkingspeed. The handheld portable environment shown in FIG. 90 can beclassified into the fixed environment and the pedestrian environment.

On the other hand, the handheld mobile environment must consider notonly the walking speed of a receiving user, but also the moving speed ofa vehicle, a train, etc. such that the handheld mobile environment cansupport the high Doppler environment.

A right column of the Table shown in FIG. 90 shows the networkconfiguration for each reception mode.

The network configuration may indicate the network structure. Thenetwork configuration according to the embodiment can be classified intoa Multi Frequency Network (MFN) composed of a plurality of frequenciesand a Single Frequency Network (SFN) composed of a single frequencyaccording to a frequency management method within the network.

MFN may indicate a network structure for transmitting a broadcast signalusing many frequencies in a wide region. A plurality of transmissiontowers located at the same region or a plurality of broadcast signaltransmitters may transmit the broadcast signal through differentfrequencies. In this case, the delay spread caused by a natural echo maybe formed by a topography, geographic features, etc. In addition, thebroadcast signal receiver is designed to receive only one radio wave,such that the reception quality can be determined according to themagnitude of a received radio wave.

SFN may indicate a network structure in which a plurality of broadcastsignal transmitters located at the same region can transmit the samebroadcast signal through the same frequency. In this case, the maximumdelay spread of a transmission (Tx) channel becomes longer due to theadditional man-made echo. In addition, the reception (Tx) quality may beaffected not only by a mutual ratio between a radio wave to be receivedand a radio wave of the jamming frequency, but also by a delay time,etc.

When deciding the Tx parameters, the guard interval value may be decidedin consideration of the maximum delay spread of the Tx channel so as tominimize the inter symbol interference. The guard interval may be aredundant data additionally inserted into the transmitted broadcastsignal, such that it is necessary to design the entire symbol durationto minimize the loss of SNR in consideration of the entire Tx powerefficiency.

FIG. 91 shows a bandwidth of the broadcast signal according to anembodiment of the present invention.

Referring to FIG. 91, the bandwidth of the broadcast signal is identicalto a waveform transform bandwidth, the waveform transform bandwidth mayinclude a channel bandwidth and a spectrum mask, and the channelbandwidth may include a signal bandwidth. The transmission (Tx)parameters according to the embodiment need to satisfy the spectrum maskrequested for minimizing interference of a contiguous channel within thecorresponding channel bandwidth allocated to the next generationbroadcast Tx/Rx system, and need to be designed for maximizing the Txefficiency within the bandwidth of the corresponding broadcast signal.In addition, a plurality of carriers can be used when theabove-mentioned waveform generation module 1300 converts input signals,the Tx parameters may coordinate or adjust the spacing among subcarriersaccording to the number of subcarriers used in the waveform transformbandwidth, the length of an entire symbol in a time domain is decided,and a transmission (Tx) mode appropriate for the Rx scenario of the nextgeneration broadcast Tx/Rx system is classified, such that the Txparameters can be designed according to the Rx scenario.

FIG. 92 shows tables including Tx parameters according to theembodiment.

FIG. 92(A) is a Table that shows guard interval values to be used as Txparameters according to the above-mentioned reception mode and thenetwork configuration. FIG. 92(B) is a Table that shows vehicle speedvalues to be used as Tx parameters according to the above-mentionedreception mode and the network configuration.

As described above, the guard interval may be designed in considerationof the maximum delay spread based on the network configuration and theRx antenna environment according to the reception (Rx) scenario.

The vehicle speed used as the Tx parameter may be designed and decidedin consideration of the network configuration and the Rx antennaenvironment according to Rx scenario categories types.

In order to implement the optimal design of the next generationbroadcast Tx/Rx system, the present invention provides a method forestablishing the guard interval (or elementary guard interval) and thevehicle speed, and optimizing Tx parameters using the optimizationscaling factor.

Symbols (or OFDM symbols) contained in the signal frame according to theembodiment may be transmitted for a specific duration. In addition, eachsymbol may include not only a guard interval region corresponding to theuseful part corresponding to the active symbol duration length, but alsothe guard interval. In this case, the guard interval region may belocated ahead of the useful part.

As shown in FIG. 92(A), the guard interval according to the embodimentmay be set to N_(G) _(_) _(a1), N_(G) _(_) _(a2,) . . . , N_(G) _(_)_(b1), N_(G) _(_) _(b2,) . . . , N_(G) _(_) _(c1), N_(G) _(_) _(c2,) . .. , N_(G) _(_) _(d1), N_(G) _(_) _(d2,) . . . , N_(G) _(_) _(e1), N_(G)_(_) _(e2,) . . . , N_(G) _(_) _(f1), N_(G) _(_) _(f2,) . . . N_(G) _(_)_(g1), N_(G) _(_) _(g2,) . . . , N_(G) _(_) _(h1), N_(G) _(_) _(h2,) . .. according to the above-mentioned reception modes.

The guard intervals (a) and (b) shown in FIG. 92(A) may show exemplaryguard intervals applicable to the next generation broadcast Tx/Rxsystem. In more detail, the guard interval (a) shows one embodiment inwhich the elementary guard interval is set to 25ρs, and the guardinterval (b) shows another embodiment in which the elementary guardinterval is set to 30 μs. In the above-mentioned embodiments, theoptimization scaling factor for implementing optimization based on anetwork structure while simultaneously optimizing Tx efficiency of Txsignals and SNR damage is set to L_(alpha1), L_(alpha2), L_(beta1), orL_(beta2).

As shown in FIG. 92(B), the vehicle speed according to the embodimentmay be set to quasi static, <V_(p) _(_) _(a1) km/h, <V_(p) _(_) _(b1)km/h, V_(m) _(_) _(a1) km/h˜V_(m) _(_) _(a2) km/h, or V_(m) _(_) _(b1)km/h˜V_(m) _(_) _(b2) km/h according to the above-mentioned receptionmodes.

The vehicle speed (a) shown in FIG. 92(B) shows an example of thevehicle speed applicable to the next generation broadcast Tx/Rx systemaccording to the embodiment.

In accordance with this embodiment, the elementary vehicle speed may beset to ‘quasi-static’, ‘3 km/h’, and ‘3 km/h˜200 km/h’ according to therespective reception scenarios, and the optimization scaling factor forimplementing optimization based on the network structure and optimizingTx efficiency of Tx signals and time-variant channel estimation may beset to V_(alpha1), V_(alpha2), V_(beta1), and V_(beta1).

The following equation may be used to decide an effective signalbandwidth (hereinafter referred to as eBW) of the optimized Tx signalsaccording to the present invention

eBW={N _(waveform) _(_) _(scaling)×(N _(pilotdensity) ×N _(eBW)+α}×Fs(Hz)  [Math FIG. 20].

In Math FIG. 20, N_(waveform) _(_) _(scaling) may denote a waveformscaling factor, N_(pilotdensity) may denote a pilot density scalingfactor, N_(eBW) may denote an effective signal bandwidth scaling factor,and a may denote an additional bandwidth factor. In addition, Fs maydenote a sampling frequency.

In order to decide the effective signal bandwidth (eBW) optimized for aspectrum mask based on a channel bandwidth, the present invention mayuse the above-mentioned factors as the optimization parameters (oroptimum parameters). Specifically, according to the equation of thepresent invention, Tx efficiency of Tx parameters can be maximized bycoordinating the waveform transform bandwidth (sampling frequency). Theindividual factors shown in Equation will hereinafter be described indetail.

The waveform scaling factor is a scaling value depending upon abandwidth of a carrier to be used for waveform transform. The waveformscaling factor according to the embodiment may be set to an arbitraryvalue proportional to the length of nonequispaced fast Fourier transform(NFFT) in case of OFDM.

The pilot density scaling factor may be established according to apredetermined position of a reference signal inserted by a referencesignal insertion and PAPR reduction block 7100, and may be establishedby the density of the reference signal.

The effective signal bandwidth scaling factor may be set to an arbitraryvalue that can satisfy a specification of a spectrum mask contained inthe Tx channel bandwidth and at the same time can maximize the bandwidthof the Tx signals. As a result, the optimum eBW can be designed.

The additional bandwidth factor may be set to an arbitrary value forcoordinating additional information and structures needed for the Txsignal bandwidth. In addition, the additional bandwidth factor may beused to improve the edge channel estimation throughput of spectrumsthrough reference signal insertion.

Number of Carrier (NoC) may be a total number of carriers transmittedthrough the signal bandwidth, and may be denoted by an equationcontained in a brace of the equation.

The broadcast signal transmission apparatus according to the presentinvention may use Tx parameters that are capable of optimizing theeffective signal bandwidth (eBW) according to the number of subcarriersused for transform. In addition, the broadcast signal transmissionapparatus according to the present invention can use the above-mentionedeffective signal bandwidth scaling factor as a transmission (Tx)parameter capable of optimizing the effective signal bandwidth (eBW).

The effective signal bandwidth (eBW) scaling factor is extended in unitsof a pilot density of a predetermined reference signal, such that theeBW scaling factor may be set to a maximum value optimized for thespectrum mask. In this case, the broadcast signal transmission apparatusaccording to the present invention coordinates the waveform transformbandwidth (i.e., sampling frequency) of vague parts capable of beinggenerated according to the pilot density unit, such that the eBW scalingfactor for the spectrum mask can be decided.

FIG. 93 shows a table including Tx parameters capable of optimizing theeffective signal bandwidth (eBW) according to the embodiment.

The Tx parameters shown in FIG. 93 can satisfy the FederalCommunications Commission (FCC) spectrum mask for the 6 MHz channelbandwidth, and can optimize the effective signal bandwidth (eBW) of thenext generation broadcast system based on the OFDM scheme.

FIG. 93(A) shows Tx parameters (See Example A) established with respectto the guard interval (a) and the vehicle speed (a). FIG. 93(B) shows Txparameters (See Example B) established with respect o the guard interval(b) and the vehicle speed (b).

FIG. 93(A′) shows a table indicating an embodiment of a GI duration forcombination of FFT and GI modes established by the concept of FIG.93(A). FIG. 93(8′) shows a table indicating an embodiment of a GIduration for combination of FFT (NFFT) and GI modes established by theconcept of FIG. 93(B).

Although the Tx parameters shown in FIGS. 93(A) and 93(B) areestablished for three FFT modes (i.e., 8K, 16K and 32K FFT modes), itshould be noted that the above Tx parameters can also be applied toother FFT modes (i.e., 1K/2K/4K/64K FFT modes) as necessary. Inaddition, FIG. 93(A) and FIG. 93(B) show various embodiments of theoptimization scaling factors applicable to the respective FFT modes.

The broadcast signal transmission apparatus according to the embodimentcan insert the reference signal into the time and frequency domains inconsideration of the Tx parameters shown in (A) and (B), the receptionscenario, and the network configuration, and the reference signal can beused as additional information for synchronization and channelestimation.

The broadcast signal transmission apparatus according to the embodimentmay establish the density (Npilotdensity) of a reference signal and theoptimized eBW in consideration the ratio of a channel estimation rangeof the guard interval. In addition, the waveform scaling factoraccording to the embodiment may be determined in proportion to the FFTsize for each FFT mode.

If a total number of the remaining carriers other than a null carrierused as a guard band during IFFT is decided by the waveform transformscheme, the broadcast signal transmission apparatus according to theembodiment may coordinate the waveform transform bandwidth (i.e.,sampling frequency) so as to determine a maximum signal bandwidth notexceeding the spectrum mask. The sampling frequency may decide theoptimized signal bandwidth, and may be sued to decide the OFDM symbolduration and the subcarrier spacing. Accordingly, the sampling frequencymay be determined in consideration of not only the guard interval, a Txchannel of the vehicle speed, and the reception scenario, but also theTx signal efficiency and the SNR damage. In FIG. 93, (A) shows anembodiment in which ‘Fs’ is set to 221/32 MHz, and (B) shows anembodiment in which ‘Fs’ is set to (1753/256) MHz.

‘fc’ in FIGS. 93(A) and 93(B) may denote the center frequency of the RFsignal, and ‘Tu’ may denote an active symbol duration.

FIG. 94 shows a table including Tx parameters for optimizing theeffective signal bandwidth (eBW) according to another embodiment of thepresent invention.

FIG. 94(A) shows a table indicating the same Tx parameters (See ExampleA) as in FIG. 93(A). FIG. 94(B) shows another embodiment of the Table ofFIG. 93(B). Table of FIG. 94(B) shows Tx parameters (See Example B-1)established with respect to the guard interval (b) and the vehicle speed(b).

FIG. 94(A′) shows a table indicating an embodiment of a GI duration forcombination of FFT and GI modes established by the concept of FIG.94(A). FIG. 94(B′) shows a table indicating an embodiment of a GIduration for combination of FFT and GI modes established by the conceptof FIG. 94(B).

Although the Tu value of the center column of FIG. 94(B) is changed to2392.6 differently from the concept of FIG. 93(B), the remainingfunctions and values of the respective Tx parameters shown in FIG. 94are identical to those of FIG. 93, and as such a detailed descriptionthereof will herein be omitted for convenience of description.

FIG. 95 shows a Table including Tx parameters for optimizing theeffective signal bandwidth (eBW) according to another embodiment of thepresent invention.

FIG. 95(A) shows a Table indicating another embodiment of the concept ofFIG. 94(B). In more detail, FIG. 95(A) is a Table including Txparameters (See Example B-2) in case that ‘Fs’ is set to 219/32 MHz.FIG. 95(B) shows a Table indicating an embodiment of a GI duration forcombination of FFT and GI modes established by the concept of FIG.95(A).

Tx parameters shown in FIG. 95(A) has a lower eBW value whereas theyhave higher values of fc and Tu, differently from the Tx parametersshown in FIG. 94(B). In this case, according to one embodiment of thepresent invention, the eBW value may be set to a specific value that iscapable of being established as a factor with respect to the channelbandwidth.

FIG. 96 shows Tx parameters according to another embodiment of thepresent invention.

As can be seen from FIG. 96(A), when establishing the scaling factor andthe Fs value corresponding to a channel bandwidth of 5, 7, or 8 MHz, theresultant scaling factor can be obtained by the product (multiplication)of a scaling factor having been calculated on the basis of the 6 MHz Fsvalue. The scaling factor may correspond to the rate of the channelbandwidth.

FIG. 96(B) is a Table including Tx parameters capable of optimizing theeffective signal bandwidth (eBW) shown in FIGS. 93 to 95.

In more detail, a Table located at an upper part of FIG. 96(B) shows Txparameters corresponding to the 5, 6, 7, 8 MHz channel bandwidths ofFIGS. 93(A) and 94(B).

The table located at the center part of FIG. 96(B) shows Tx parameterscorresponding to the 5, 6, 7, 8 MHz channel bandwidths of the example(B-1) of FIG. 94.

The table located at the lower part of FIG. 96(B) shows Tx parameterscorresponding to the channel bandwidth shown in the example (B-2) ofFIG. 95.

Referring to the second row of FIG. 96(A), the Fs value corresponding toeach channel bandwidth in the upper end of FIG. 96(B) is calculated bythe product of the scaling factor having been calculated on the basis ofthe 6 MHz Fs value.

Referring to the third row of FIG. 96(A), the Fs value corresponding toeach channel bandwidth in the center part of FIG. 96(B) is calculated bythe product of the scaling factor having been calculated on the basis ofthe 6 MHz Fs value. Referring to the third row of FIG. 96(A), the Fsvalue corresponding to each channel bandwidth in the lower part of FIG.96(B) is calculated by the product of the scaling factor having beencalculated on the basis of the 6 MHz Fs value.

FIG. 97 is a graph indicating Power Spectral Density (PSD) of atransmission (Tx) signal according to an embodiment of the presentinvention.

FIG. 97 shows the Power Spectral Density (PSD) calculated using theabove-mentioned Tx parameters when the channel bandwidth is set to 6MHz.

The left graph of FIG. 97(A) shows the PSD of the Tx signal optimizedfor the FCC spectrum mask of the example (A) of FIGS. 93 and 94. Theright graph of FIG. 97(A) shows the enlarged result of some parts of theleft graph.

The left graph of FIG. 97(B) shows the PSD of the Tx signal optimizedfor the FCC spectrum mask of the example (B) of FIG. 93. The right graphof FIG. 97(B) shows the enlarged result of some parts of the left graph.

As shown in the right graph of (A) and (B), individual graphs show notonly lines for designating the FCC spectrum mask specification, but alsolines indicating PSD of the Tx signal derived using Tx parameterscorresponding to 8K, 16K and 32K.

In order to optimize the Tx signal efficiency as shown in FIG. 97, thePSD of each Tx signal need not exceed a threshold value of the spectrummask at a breakpoint of the target spectrum mask. In addition, a band ofthe PSD of an out-of-band emission Tx signal may be limited by abaseband filter as necessary.

FIG. 98 is a table showing information related to the reception modeaccording to another embodiment of the present invention.

FIG. 98 shows another embodiment of the Table showing informationrelated to the reception mode of FIG. 90. Table of FIG. 98 shows anetwork configuration, an FFT value (NFFT), a guard interval, and avehicle speed, that correspond to each reception mode. The guardinterval and the vehicle speed of FIG. 98 are identical to those of FIG.92.

Since the fixed rooftop environment corresponds to a time-variant Txchannel environment, it is less affected by Doppler, such that alarge-sized FFT such as 16K, 32K, etc. can be used. In addition, datatransmission can be carried out in a manner that a higher data Txefficiency can be achieved in the redundancy ratio such as the guardinterval, the reference signal, etc. appropriate for the networkconfiguration.

In case of the handheld portable environment, a low-level Dopplerenvironment can be supported as the indoor/outdoor receptionenvironments in consideration of mobility such as an adult walkingspeed, and FFT such as 8K, 16K, 32K, etc. capable of supporting a highfrequency sensitivity can be used.

The handheld mobile environment must consider not only the walking speedof a receiving user, but also the moving speed of a vehicle, a train,etc. such that the handheld mobile environment can support the highDoppler environment, and can use 4K-, 8K-, and 16K-FFT capable ofsupporting a relatively low frequency sensitivity.

The guard interval according to an embodiment of the present inventionmay be established to support the same-level coverage in considerationof the network configuration for each reception.

The following description proposes the pilot pattern used as a referencesignal for Tx channel estimation and the pilot mode for the same Txchannel estimation on the basis of the above embodiments of theabove-mentioned Tx parameters.

The broadcast signal transmission apparatus or the above-mentionedwaveform transform block 7200 according to the embodiment can insert aplurality of pilots into a signal frame generated from the framestructure module 1200, and can OFDM-modulate the broadcast signals usingthe Tx parameters. Various cells contained in the OFDM symbol may bemodulated using reference information (i.e., pilots). In this case, thepilots may be used to transmit information known to the broadcast signalreceiver, and the individual pilots may be transmitted at a power levelspecified by a pilot pattern.

The pilots according to the embodiment of the present invention may beused for frame synchronization, frequency and time synchronization,channel estimation, etc.

The pilot mode according to the embodiment of the present invention maybe specific information for indicating pilots which reduce overhead ofTx parameters and are established to transmit the optimized broadcastsignal. The above-mentioned pilot pattern and pilot mode may equally beapplied to the above-mentioned reception mode and network configuration.In addition, the pilot pattern and pilot mode according to theembodiment can be applied to data symbols contained in the signal frame.

FIG. 99 shows the relationship between a maximum channel estimationrange and a guard interval according to the embodiment.

As described above, Math FIG. 20 is used to decide the effective signalbandwidth (eBW) of the Tx signal, and may use the pilot density scalingfactor as an optimization parameter. In this case, Math FIG. 20 may bedecided by optimizing time- and frequency-arrangement of the pilotsignal for SISO channel estimation, a pilot density related to dataefficiency, and Dx and Dy values.

The pilot density may correspond to the product of a distance betweenpilots of the time and frequency domains, and pilot overhead occupied bypilots of the symbol may correspond to an inverse number of the pilotdensity.

Dx may denote a distance between pilots in a frequency domain, and Dymay denote a distance between pilots in a time domain. Dy may be used todecide the maximum tolerable Doppler speed. Accordingly, Dy may be setto a specific value that is optimized in consideration of the vehiclespeed decided according to Rx scenario categories.

As described above, the pilot density may be used to decide the pilotoverhead, and the Dx and Dy values may be decided in consideration ofthe Tx channel state and the Tx efficiency.

The maximum channel estimation range (TChEst) shown in FIG. 99 may bedecided by dividing the Tx parameter (Tu) by the Dx value.

The guard interval having a predetermined length, the pre-echo region,and the post-echo region may be contained in the maximum channelestimation range.

The ratio of a given guard interval and a maximum channel estimationrange may indicate a margin having a channel estimation range forestimating the guard interval. If the margin value of the channelestimation range exceeds the guard interval length, values exceeding theguard interval length may be assigned to the pre-echo region and thepost-echo region. The pre-echo region and the post-echo region may beused to estimate the channel impulse response exceeding the guardinterval length, and may be used as a region to be used for estimationand compensation of a timing error generable in a synchronizationprocess. However, if the margin is increased in size, the pilot overheadis unavoidably increased so that Tx efficiency can be reduced.

FIGS. 100 and 101 show Tables in which pilot parameters depending on theguard intervals (A) and (B) and the vehicle speed are defined, and thetables shown in FIGS. 100 and 101 will hereinafter be described indetail.

FIG. 100 shows a Table in which pilot parameters are defined accordingto an embodiment of the present invention.

FIG. 100 shows the pilot parameters according to the guard interval (A)and the vehicle speed. FIG. 100(A) is a table indicating pilot patternsfor use in the SISO and MIXO Tx channels, FIG. 100(B) shows theconfiguration of a pilot pattern for use in the SISO and MIXO Txchannels, and FIG. 100(C) is a table indicating the configuration of apilot pattern for use in the MIXO Tx channel.

In more detail, FIG. 100(A) shows the pilot pattern decided for eachpilot density value and the Dx and Dy values defined in each of the SISOand MIXO Tx channels. The pilot pattern according to this embodiment maybe denoted by PP5-4 in which a first number denotes the Dx value and asecond number denotes the Dy value. If the Dx value in the same pilotdensity is reduced, the pilot pattern can support a longer delay spread.If the Dy value is reduced, the pilot pattern can adaptively cope with afaster Doppler environment.

FIG. 100(B) and FIG. 100(C) show Tables including the guard intervalduration and the pilot pattern configuration depending on the FFT value.In more detail, numbers shown in the first row of each table shown in(B) and (C) may denote the guard interval duration. The first column maydenote FFT (NFFT) values described in FIGS. 93 to 96. However, althoughFIGS. 100(B) and 100(C) equally show the configuration of the pilorpattern for use in the MIXO case, there is a difference in FIGS. 100(B)and 100(C) in that FIG. 100(B) shows the MIXO-1 pilot pattern having alarger pilot overhead, and FIG. 100(C) shows the MIXO-2 pilot patternhaving a lower mobility.

The duration of the guard interval shown in FIGS. 100(B) and 100(C) isconceptually identical to the guard interval length shown in FIG. 99. Inaccordance with the embodiment of the present invention, 25 μs, 50 μs,100 μs, 200 μs, and 400 μs values may be used in consideration of themaximum delay spread, and the FFT size may be set to 8K, 16K and 32K.

As can be seen from (A), the Dx value may be set to 5, 10, 20, 40, 80,or 160 in consideration of the guard interval duration and the FFT size.In this case, an elementary Dx value (5) acting as a basic value may bedefined as a changeable value depending on each Tx mode, and may beestablished in consideration of about 20% of the margin value of theabove-mentioned channel estimation range. In addition, according to oneembodiment of the present invention, the margin value of the channelestimation range may be coordinated or adjusted using the L_(alpha1)value in MFN and using the L_(alpha2) value in SFN as shown in FIGS.92(A) and 92 (B).

The Dy value may be established according to a reception (Rx) scenarioand the Tx mode dependent upon the Rx scenario. Accordingly, the Dyvalue may be assigned different values according to the SISO or MIXO Txchannel. As shown in the drawing, Dy may be set to 2, 4 or 8 in case ofthe SISO Tx channel according to an embodiment of the present invention.

The MIXO Tx channel is classified into the MIXO-1 version having largepilot overhead and the MIXO-2 version having lower mobility, such thatthe Dy value can be established in different ways according toindividual versions.

The MIXO-1 version having large overhead increases the pilot overhead,so that l can support the same maximum delay spread and the same maximummobile speed in the same network configuration as in the SISO Txchannel. In this case, the Dy value may be set to 2, 4 or 8 in the samemanner as in the SISO Tx channel. That is, the MIXO-1 Tx channel can beapplied not only to the above-mentioned handheld portable environmentbut also the handheld mobile environment.

The MIXO-2 version having low mobility is designed to guarantee the samecoverage and capacity as in the SISO Tx channel although the MIXO-2version has a little damage in terms of the mobile speed support. Inthis case, the Dy value may be set to 4, 8, or 16.

FIG. 101 shows a Table in which pilot parameters of another embodimentare defined.

In more detail, FIG. 101 shows the pilot parameters according to theguard interval (B) and the vehicle speed. FIG. 101(A) is a tableindicating pilot patterns for use in the SISO and MIXO Tx channels, FIG.101(B) shows the configuration of a pilot pattern for use in the SISOand MIXO Tx channels, and FIG. 101(C) is a table indicating theconfiguration of a pilot pattern for use in the MIXO Tx channel.

Functions and contents of the pilot parameters shown in FIG. 101 areidentical to those of FIG. 100, and as such a detailed descriptionthereof will herein be omitted for convenience of description.

The structure and location of pilots for MIXO (MISO, MIMO) Tx channelestimation may be established through the above-mentioned pilotpatterns. The nulling encoding and the Hadamard encoding scheme may beused as the pilot encoding scheme for isolating each Tx channelaccording to one embodiment of the present invention.

The following Math FIG. 21 may be used to indicate the nulling encodingscheme.

$\begin{matrix}{\begin{bmatrix}y_{{tx}\; 1} \\y_{{tx}\; 2}\end{bmatrix} = {\begin{bmatrix}1 & 0 \\0 & 1\end{bmatrix}\begin{bmatrix}p_{{tx}\; 1} \\p_{{tx}\; 2}\end{bmatrix}}} & \left\lbrack {{Math}\mspace{14mu} {Figure}\mspace{14mu} 21} \right\rbrack\end{matrix}$

The nulling encoding scheme has no channel interference in estimatingrespective channels, the channel estimation error can be minimized, andan independent channel can be easily estimated in the case of usingsymbol timing synchronization. However, since the pilot gain must beamplified to derive a channel estimation gain, the influence of InterChannel Interference (ICI) of contiguous data caused by the pilot basedon a time-variant channel is relatively high. In addition, if the pilotsto be allocated to individual channels according to the pilotarrangement have different locations, the SNR of effective data may bechanged per symbol. The MIXO-1 pilot pattern according to theabove-mentioned embodiment may also be effectively used even in thenulling encoding scheme, and a detailed description thereof willhereinafter be described in detail.

The following equation may be used to indicate the nulling encodingscheme.

$\begin{matrix}{\begin{bmatrix}y_{{tx}\; 1} \\y_{{tx}\; 2}\end{bmatrix} = {\begin{bmatrix}1 & 1 \\1 & {- 1}\end{bmatrix}\begin{bmatrix}p_{{tx}\; 1} \\p_{{tx}\; 2}\end{bmatrix}}} & \left\lbrack {{Math}\mspace{14mu} {Figure}\mspace{14mu} 22} \right\rbrack\end{matrix}$

In case of the Hadamard encoding scheme, the Hadamard encoding schemecan perform channel estimation through simple linear calculation, andcan obtain a gain caused by the noise average effect as compared to thenulling encoding scheme. However, the channel estimation errorencountered in the process for obtaining an independent channel mayunexpectedly affect other channels, and there may occur ambiguity in thesymbol timing synchronization using pilots.

The broadcast signal transmission apparatus according to the embodimentof the present invention may establish the above-mentioned two encodingschemes described as the MIXO pilot encoding scheme according to thereception (Rx) scenario and the Tx channel condition in response to apredetermined mode. The broadcast signal reception apparatus accordingto the embodiment may perform channel estimation through a predeterminedmode.

FIG. 102 shows the SISO pilot pattern according to an embodiment of thepresent invention.

The pilot pattern shown in FIG. 102 indicates the SISO pilot pattern foruse in the case in which the pilot density of FIG. 101 is set to 32.

As described above, the pilots may be inserted into a data symbol regionof the signal frame. In FIG. 102, a horizontal axis of the pilot patternmay denote a frequency axis, and a vertical axis thereof may denote atime axis. In addition, pilots successively arranged at both ends of thepilot pattern may indicate reference signals that are inserted tocompensate for distortion at the edge of a spectrum generated by channelestimation.

In more detail, FIG. 102(A) shows an exemplary pilot pattern denoted byPP4-8, FIG. 102(B) shows an exemplary pilot pattern denoted by PP8-4,and FIG. 102(C) shows an exemplary pilot pattern denoted by PP16-2. Inother words, as can be seen from FIG. 102(A), pilots may be periodicallyinput in units of 4 carriers on the frequency axis, and each pilot maybe input in units of 8 symbols on the time axis. FIG. 102(B) and FIG.102(C) may also illustrate the pilot patterns having been input in thesame manner.

The pilot pattern of another pilot density shown in FIG. 101 may bedenoted by coordination of the Dx and Dy values.

FIG. 103 shows the MIXO-1 pilot pattern according to an embodiment ofthe present invention.

The pilot pattern of FIG. 103 shows the MIXO-1 pilot pattern for use inthe case that the pilot density of FIG. 101 is set to 32. The pilotpattern of FIG. 103 is used in the case that two Tx antennas exist.

As described above, a horizontal axis of the pilot pattern may denote afrequency axis, and a vertical axis of the pilot pattern may denote atime axis. The pilots successively arranged at both edges of the pilotpattern may be reference signals that have been inserted to compensatefor distortion at a spectrum edge encountered in the channel estimationprocess.

In more detail, (A) may denote an exemplary case in which the pilotpattern is denoted by PP4-8, (B) may denote an exemplary case in whichthe pilot pattern is denoted by PP8-4, and (C) may denote an exemplarycase in which the pilot pattern is denoted by PP16-2.

In order to discriminate among the individual MIXO Tx channels, pilotstransmitted to the respective Tx channels may be arranged contiguous toeach other in the frequency domain according to an embodiment of thepresent invention. In this case, the number of pilots allocated to twoTx channels within one OFDM symbol is set to the same number.

As shown in the drawing, the MIXO-1 pilot pattern according to anembodiment has an advantage in that a data signal is arranged at thenext position of a channel estimation pilot even when a reference signalfor synchronization estimation is arranged, so that correlation betweensignals is reduced at the same carrier and the synchronizationestimation throughput is not affected by the reduced correlation.

In case of the MIXO-1 pilot pattern according to an embodiment, evenwhen the broadcast signal transmission apparatus performs pilot encodingusing the above-mentioned nulling encoding scheme, broadcast signalshaving the same Tx power can be transmitted to the individual Txantennas, such that the broadcast signals can be transmitted withoutadditional devices or modules for compensating for variation of Txsignals. That is, in case of using the MIXO-1 pilot pattern according toan embodiment, the MIXO-1 pilot pattern is not affected by the pilotencoding scheme, and pilot power is coordinated by the pilot encodingscheme, such that the channel estimation throughput of the broadcastsignal reception apparatus can be maximized.

The pilot pattern of another pilot density shown in FIG. 101 may bedenoted by coordination of the Dx and Dy values.

FIG. 104 shows the MIXO-2 pilot pattern according to an embodiment ofthe present invention.

The pilot pattern of FIG. 104 shows the MIXO-2 pilot pattern for use inthe case that the pilot density of FIG. 101 is set to 32. The pilotpattern of FIG. 104 is used in the case that two Tx antennas exist.

As described above, a horizontal axis of the pilot pattern may denote afrequency axis, and a vertical axis of the pilot pattern may denote atime axis. The pilots successively arranged at both edges of the pilotpattern may be reference signals that have been inserted to compensatefor distortion at a spectrum edge encountered in the channel estimationprocess.

In more detail, (A) may denote an exemplary case in which the pilotpattern is denoted by PP4-16, (B) may denote an exemplary case in whichthe pilot pattern is denoted by PP8-8, and (C) may denote an exemplarycase in which the pilot pattern is denoted by PP16-4.

As described above, the MIXO-2 pilot pattern is designed to cut thesupported mobility in half, instead of supporting the same capacity, thesame pilot overhead, and the same coverage as those of the SISO Txchannel.

Tx channels are semi-statically used in the reception scenario in whichthe UHDTV service must be supported so that the serious problem does notoccur. The MIXO-2 pilot pattern according to an embodiment can be usedto maximize the data Tx efficiency in the reception scenario in whichthe UHDTV service must be supported.

The pilot pattern of another pilot density shown in FIG. 101 may bedenoted by coordination of the Dx and Dy values.

FIG. 105 illustrates a MIMO encoding block diagram according to anembodiment of the present invention.

The MIMO encoding scheme according to an embodiment of the presentinvention is optimized for broadcasting signal transmission. The MIMOtechnology is a promising way to get a capacity increase but it dependson channel characteristics. Especially for broadcasting, the strong LOScomponent of the channel or a difference in the received signal powerbetween two antennas caused by different signal propagationcharacteristics can make it difficult to get capacity gain from MIMO.The MIMO encoding scheme according to an embodiment of the presentinvention overcomes this problem using a rotation-based pre-coding andphase randomization of one of the MIMO output signals. MIMO encoding canbe intended for a 2×2 MIMO system requiring at least two antennas atboth the transmitter and the receiver.

MIMO processing can be required for the advanced profile frame, whichmeans all DPs in the advanced profile frame are processed by the MIMOencoder (or MIMO encoding module). MIMO processing can be applied at DPlevel. Pairs of the Constellation Mapper outputs NUQ (e_(1,i) ande_(2,i)) can be fed to the input of the MIMO Encoder. Paired MIMOEncoder output (g_(1,i), and g_(2,i)) can be transmitted by the samecarrier k and OFDM symbol l of their respective TX antennas.

The illustrated diagram shows the MIMO Encoding block, where i is theindex, of the cell pair of the same XFECBLOCK and Ncells is the numberof cells per one XFECBLOCK.

FIG. 106 shows a MIMO encoding scheme according to one embodiment of thepresent invention.

If MIMO is used, a broadcast/communication system may transmit moredata. However, channel capacity of MIMO may be changed according tochannel environment. In addition, if Tx and Rx antennas are different interms of power or if correlation between channel is high, MIMOperformance may deteriorate.

If dual polar MIMO is used, two components may reach a receiver atdifferent power ratios according to propagation property ofvertical/horizontal polarity. That is, if dual polar MIMO is used, powerimbalance may occur between vertical and horizontal antennas. Here, dualpolar MIMO may mean MIMO using vertical/horizontal polarity of anantenna.

In addition, correlation between channel components may increase due toLOS environment between Tx and Rx antennas.

The present invention proposes a MIMO encoding/decoding technique forsolving problems occurring upon using MIMO, that is, a techniquesuitable for a correlated channel environment or a power imbalancedchannel environment. Here, the correlated channel environment may be anenvironment in which channel capacity is lowered and system operation isinterrupted if MIMO is used.

In particular, in a MIMO encoding scheme, a PH-eSM PI method and afull-rate full-diversity (FRFD) PH-eSM PI method are proposed inaddition to an existing PH-eSM method. The proposed methods may be MIMOencoding methods considering complexity of a receiver and a powerimbalanced channel environment. These two MIMO encoding schemes have norestriction on the antenna polarity configuration.

The PH-eSM PI method can provide capacity increase with relatively lowcomplexity increase at the receiver side. The PH-eSM PI method may bereferred to as a full-rate spatial multiplexing (FR-SM), FR-SM method, aFR-SM encoding process, etc. In the PH-eSM PI method, rotation angle isoptimized to overcome power imbalance with complexity of O (M2). In thePH-eSM PI method, it is possible to effectively cope with spatial powerimbalance between Tx antennas.

The FRFD PH-eSM PI method can provide capacity increase and additionaldiversity gain with a relatively great complexity increase at thereceiver side. The FRFD PH-eSM PI method may be referred to as afull-rate full-diversity spatial multiplexing (FRFD-SM), an FRFD-SMmethod, FRFD-SM encoding process, etc. In the FRFD PH-eSM PI method,additional Frequency diversity gain is achieved by adding complexity ofO (M4). In the FRFD PH-eSM PI method, unlike the PH-eSM PI method, it ispossible to effectively cope not only with power imbalance between Txantennas and but also with power imbalance between carriers.

In addition, the PH-eSM PI method and the FRFD PH-eSM PI method may beMIMO encoding schemes applied to symbols mapped to non-uniform QAM,respectively. Here, mapping to non-uniform QAM may mean thatconstellation mapping is performed using non-uniform QAM. Non-uniformQAM may be referred to as NU QAM, NUQ, etc. PH-eSM PI method and FRFDPH-eSM PI method can also be applied to symbols mapped onto either QAM(uniform QAM) or Non-uniform constellation. The MIMO encoding schemeapplied to symbols mapped to non-uniform QAM may have better BERperformance than the MIMO encoding scheme applied to symbols mapped toQAM (uniform QAM) per code rate in a power imbalanced situation.However, with certain code rate and bit per channel use, applying MIMOencoding to symbols mapped onto QAM performs better.

In addition, the PH-eSM method may also be applied to non-uniform QAM.Therefore, the present invention further proposes a PH-eSM methodapplied to symbols mapped to non-uniform QAM.

Hereinafter, constellation mapping will be described.

In constellation mapper, each cell word (c_(0,i), c_(1,i), . . . ,c_(η mod−1,i)) from the Bit Interleaver in the base and the handheldprofiles, or cell word (d_(i,0,l), d_(i,1,l), . . . , d_(i,η mod−1,l),where 1=1, 2) from the Cell-word Demultiplexer in the advanced profilecan be modulated using either QPSK, QAM-16, non-uniform QAM (NUQ-64,NUQ-256, NUQ-1024) or non-uniform constellation (NUC-16, NUC-64,NUC-256, NUC-1024) to give a power-normalized constellation point, e₁.

This constellation mapping is applied only for DPs. The constellationmapping for PLS1 and PLS2 can be different.

QAM-16 and NUQs are square shaped, while NUCs have arbitrary shape. Wheneach constellation is rotated by any multiple of 90 degrees, the rotatedconstellation overlaps with its original one. This ‘rotation-sense’symmetric property makes the capacities and the average powers of thereal and imaginary components equal to each other. Both NUQs and NUCsare defined specifically for each code rate and the particular one usedis signaled by the parameter DP_MOD in PLS2. The constellation shapesfor each code rate mapped onto the complex plane will be describedbelow. Hereinafter, the PH-eSM method and the PH-eSM PI method will bedescribed. A MIMO encoding equation used for the PH-eSM method and thePH-eSM PI method is expressed as follows.

$\begin{matrix}{{\begin{bmatrix}{X_{1}\left( f_{1} \right)} \\{X_{2}\left( f_{1} \right)}\end{bmatrix} = {{{\frac{1}{\sqrt{1 + a^{2}}}\begin{bmatrix}1 & 0 \\0 & ^{j\; {\varphi {(q)}}}\end{bmatrix}}\begin{bmatrix}1 & a \\a & {- 1}\end{bmatrix}}\begin{bmatrix}S_{1} \\S_{2}\end{bmatrix}}}{or}{\underset{\underset{X}{}}{\begin{bmatrix}{X_{1}\left( f_{1} \right)} \\{X_{2}\left( f_{1} \right)}\end{bmatrix}} = {\underset{\underset{P}{}}{{\frac{1}{\sqrt{1 + a^{2}}}\begin{bmatrix}1 & 0 \\0 & ^{j\; {\varphi {(q)}}}\end{bmatrix}}\begin{bmatrix}1 & {- a} \\a & {- 1}\end{bmatrix}}\underset{\underset{S}{}}{\begin{bmatrix}S_{1} \\S_{2}\end{bmatrix}}}}} & \left\lbrack {{Math}\mspace{14mu} {figure}\mspace{14mu} 23} \right\rbrack\end{matrix}$

That is, the above equation may be expressed as X=PS. Here, S₁ and S₂may denote a pair of input symbols. Here, P may denote a MIMO encodingmatrix. Here, X₁ and X₂ may denote paired MIMO encoder outputs subjectedto MIMO encoding.

In the above equation, e^(jφ(q)) may be expressed as follows.

$\begin{matrix}{{^{{j\varphi}{(q)}} = {{\cos \; {\varphi (q)}} + {j\; \sin \; {\varphi (q)}}}},{{\varphi (q)} = {\frac{2\pi}{N}q}},{q = 0},\ldots \mspace{14mu},{N_{data} - 1},\left( {N = 9} \right)} & \left\lbrack {{Math}\mspace{14mu} {figure}\mspace{14mu} 24} \right\rbrack\end{matrix}$

According to another embodiment, the MIMO encoding equation used for thePH-eSM method and the PH-eSM PI method may be expressed as follows.

$\begin{matrix}{{\begin{bmatrix}g_{1,i} \\g_{2,i}\end{bmatrix} = {{{\frac{1}{\sqrt{1 + a^{2}}}\begin{bmatrix}1 & 0 \\0 & ^{j\; {\varphi {()}}}\end{bmatrix}}\begin{bmatrix}1 & a \\a & {- 1}\end{bmatrix}}\begin{bmatrix}e_{1,i} \\e_{2,i}\end{bmatrix}}},{{\varphi (i)} = {\frac{2\pi}{N}i}},\left( {N = 9} \right),{i = 0},\ldots \mspace{14mu},{\frac{N_{cells}}{2} - 1}} & \left\lbrack {{Math}\mspace{14mu} {figure}\mspace{14mu} 25} \right\rbrack\end{matrix}$

The PH-eSM PI method can include two steps. The first step can bemultiplying the rotation matrix with the pair of the input symbols forthe two TX antenna paths, and the second step can be applying complexphase rotation to the symbols for TX antenna 2.

The signals X₁ and X₂ to be transmitted may be generated using twotransmitted symbols (e.g., QAM symbols) S₁ and S₂. In case of atransmission and reception system using OFDM, X₁(f₁), X₂(f₂) may becarried on a frequency carrier f₁ to be transmitted. X₁ may betransmitted via a Tx antenna 1 and X₂ may be transmitted via a Txantenna 2. Accordingly, even when power imbalance is present between twoTx antennas, efficient transmission with minimum loss is possible.

At this time, if the PH-eSM method is applied to symbols mapped to QAM,a value a may be determined according to QAM order as follows. This maybe a value a when the PH-eSM method is applied to symbols mapped touniform QAM.

$\begin{matrix}{{a = {{\frac{\sqrt{2} + 2^{\frac{n}{2}}}{\sqrt{2} + 2^{\frac{n}{2}} - 2}\mspace{14mu} {for}\mspace{14mu} 2^{n}{QAM}} + {2^{n}{QAM}}}},{a = \left\{ \begin{matrix}{\sqrt{2} + 1} & {{{for}\mspace{14mu} {QPSK}} + {QPSK}} \\\frac{\sqrt{2} + 4}{\sqrt{2} + 2} & {{{for}\mspace{14mu} 16\; {QAM}} + {16\; {QAM}}} \\\frac{\sqrt{2} + 8}{\sqrt{2} + 6} & {{{for}\mspace{14mu} 64\; {QAM}} + {64\; {QAM}}} \\\frac{\sqrt{2} + 16}{\sqrt{2 + 14}} & {{{for}\mspace{14mu} 256\; {QAM}} + {256\; {QAM}}}\end{matrix} \right.}} & \left\lbrack {{Math}\mspace{14mu} {figure}\mspace{14mu} 26} \right\rbrack\end{matrix}$

At this time, if the PH-eSM PI method is applied to symbols mapped toQAM, a value a may be determined according to QAM order as follows. Thismay be a value a when the PH-eSM PI method is applied to symbols mappedto QAM (uniform QAM).

$\begin{matrix}{{a = {\sqrt{2} + {\left( {2^{\frac{n}{2}} - 1} \right)\mspace{14mu} {for}\mspace{14mu} 2^{n}{QAM}} + {2^{n}{QAM}}}},{a = \left\{ \begin{matrix}{\sqrt{2} + 1} & {{{for}\mspace{14mu} {QPSK}} + {QPSK}} \\{\sqrt{2} + 3} & {{{for}\mspace{14mu} 16\; {QAM}} + {16\; {QAM}}} \\{\sqrt{2} + 7} & {{{for}\mspace{14mu} 64\; {QAM}} + {64\; {QAM}}} \\{\sqrt{2} + 15} & {{{for}\mspace{14mu} 256\; {QAM}} + {256\; {QAM}}}\end{matrix} \right.}} & \left\lbrack {{Math}\mspace{14mu} {figure}\mspace{14mu} 27} \right\rbrack\end{matrix}$

At this time, the value a may enable a broadcast/transmission system toobtain good BER performance when considering Euclidean distance andHamming distance if X₁ and X₂ are received through a fully correlatedchannel and are decoded. In addition, the value a may enable thebroadcast/communication system to obtain good BER performance whenconsidering Euclidean distance and Hamming distance if X₁ and X₂ areindependently decoded at the receiver side (that is, if S₁ and S₂ aredecoded using X₁ and S₁ and S₂ are decoded using X₂).

The PH-eSM PI method is different from the PH-eSM method in that thevalue a is optimized in a power imbalanced situation. That is, in thePH-eSM PI method, a rotation angle value is optimized in a powerimbalance situation. In particular, when the PH-ESM PI method is appliedto symbols mapped to non-uniform QAM, the value a may be optimized ascompared to the PH-eSM method.

The above-described value a is merely exemplary and may be changedaccording to embodiment.

The receiver used for the PH-eSM method and the PH-eSM PI method maydecode a signal using the above-described MOMI encoding equation. Atthis time, the receiver may decode a signal using ML, Sub-ML (Sphere)decoding, etc.

Hereinafter, an FRFD PH-eSM PI method will be described. The MIMOencoding equation used for the FRFD PH-eSM PI method is as follows.

$\begin{matrix}{\begin{bmatrix}{X_{1}\left( f_{1} \right)} & {X_{1}\left( f_{2} \right)} \\{X_{2}\left( f_{1} \right)} & {X_{2}\left( f_{2} \right)}\end{bmatrix} = {\quad{{\frac{1}{\sqrt{1 + a^{2}}}\left. \quad{\begin{bmatrix}1 & 0 \\0 & ^{j\; {\varphi {(q)}}}\end{bmatrix}\overset{\overset{{Frequency}\mspace{14mu} {diversity}}{}}{\begin{bmatrix}{S_{1} + {aS}_{2}} & {{aS}_{3} - S_{4}} \\{S_{3} + {aS}_{4}} & {{aS}_{1} - S_{2}}\end{bmatrix}}} \right\}^{{Spatial}\mspace{14mu} {diversity}}\mspace{20mu} {{or}\begin{bmatrix}{X_{1}\left( f_{1} \right)} & {X_{1}\left( f_{2} \right)} \\{X_{2}\left( f_{1} \right)} & {X_{2}\left( f_{2} \right)}\end{bmatrix}}} = {{\frac{1}{\sqrt{1 + a^{2}}}\begin{bmatrix}1 & 0 \\0 & ^{j\; {\varphi {(q)}}}\end{bmatrix}}\begin{bmatrix}{S_{1} - {aS}_{2}} & {{aS}_{3} + S_{4}} \\{S_{3} - {aS}_{4}} & {{aS}_{1} + S_{2}}\end{bmatrix}}}}} & \left\lbrack {{Math}\mspace{14mu} {figure}\mspace{14mu} 28} \right\rbrack\end{matrix}$

By using two antennas X₁ and X₂, it is possible to obtain spatialdiversity. In addition, by utilizing two frequencies f₁ and f₂, it ispossible to obtain frequency diversity.

According to another embodiment of the present invention, a MIMOencoding scheme used for the FRFD PH-eSM PI method may be expressed asfollows.

$\begin{matrix}{\begin{bmatrix}g_{1,{2i}} & g_{1,{{2i} + 1}} \\g_{2,{2i}} & g_{2,{{2i} + 1}}\end{bmatrix} = {{\frac{1}{\sqrt{1 + a^{2}}}\begin{bmatrix}1 & 0 \\0 & ^{j\; {\varphi {()}}}\end{bmatrix}}{\quad{\begin{bmatrix}{e_{1,{2i}} + {ae}_{2,{2i}}} & {{ae}_{1,{{2i} + 1}} - e_{2,{{2i} + 1}}} \\{e_{1,{{2i} + 1}} + {ae}_{2,{{2i} + 1}}} & {{ae}_{1,{2i}} - e_{2,{2i}}}\end{bmatrix},\mspace{20mu} {{\varphi (i)} = {\frac{2\pi}{N}i}},\mspace{20mu} \left( {N = 9} \right),\mspace{20mu} {i = 0},\ldots \mspace{14mu},{\frac{N_{cells}}{4} - 1}}}}} & \left\lbrack {{Math}\mspace{14mu} {figure}\mspace{14mu} 29} \right\rbrack\end{matrix}$

The FRFD PH-eSM PI method can take two pairs of NUQ symbols (or UniformQAM symbols or NUC symbols) as input to provide two pairs of MIMO outputsymbols.

The FRFD PH-eSM PI method requires more decoding complexity of areceiver but may have better performance. According to the FRFD PH-eSMPI method, a transmitter generates signals X₁(f₁), X₂(f₁), X₁(f₂) andX₂(f₂) to be transmitted using four transmit symbols S₁, S₂, S₃, S₄. Atthis time, the value a may be equal to the value a used for theabove-described PH-eSM PI method. This may be a value a when the FRFDPH-eSM method is applied to symbols mapped to QAM (uniform QAM).

The MIMO encoding equation of the FRFD PH-eSM PI method may usefrequency carriers f₁ and f₂ unlike the MIMO encoding equation of theabove-described PH-eSM PI method. Therefore, the FRFD PH-eSM PI methodmay efficiently cope not only with power imbalance between Tx antennasbut also with power imbalance between carriers.

In association with MIMO encoding, a structure for additionallyobtaining frequency diversity may include Golden code, etc. The FRFDPH-eSM PI method according to the present invention can obtain frequencydiversity with complexity lower than that of Golden code.

FIG. 107 is a diagram showing a PAM grid of an I or Q side according tonon-uniform QAM according to one embodiment of the present invention.

The above-described PH-eSM PI and FRFD PH-eSM PI methods are applicableto symbols mapped to non-uniform QAM. Non-uniform QAM is a modulationscheme which obtains higher capacity by adjusting a PAM grid value perSNR unlike QAM (uniform QAM). It is possible to obtain more gain byapplying MIMO to symbols mapped to non-uniform QAM. In this case, theencoding equations of the PH-eSM PI and FRFD PH-eSM PI methods are notchanged but a new value “a” may be necessary when the PH-eSM PI and FRFDPH-eSM PI methods are applied to symbols mapped to non-uniform QAM. Thisnew value “a” may be obtained using the following equation.

$\begin{matrix}{{a = {{b\left( {P_{m} - P_{m - 1}} \right)} + P_{m}}}{for}{{{2^{n}{QAM}} + {2^{n}{QAM}}},{m = 2^{\frac{n}{2} - 1}}}{for}{2^{n}{QAM}}} & \left\lbrack {{Math}\mspace{14mu} {figure}\mspace{14mu} 30} \right\rbrack\end{matrix}$

This new value “a” may be a value a when the PH-eSM PI and FRFD PH-eSMPI methods are applied to symbols mapped to non-uniform QAM.

As shown in this figure, the PAM grid of the I or Q side used fornon-uniform QAM is defined and the new value “a” may be obtained using alargest value P_(m) and a second largest value P_(m−1) of this grid. Asignal transmitted via the Tx antenna may be suitably decoded using thisnew value “a” alone.

In the equation for generating the new value “a”, b denotes asub-constellation separation factor. By adjusting the value b, adistance between sub-constellations present in a MIMO encoded signal maybe adjusted. In case of non-uniform AM, since a distance betweenconstellations (or a distance between sub-constellations) is changed, avariable b may be necessary. Examples of the value b may include

$\frac{\sqrt{2}}{2}.$

This value may be obtained by Hamming distance and Euclidean distancebased on a point having highest power on a constellation and pointsadjacent thereto.

In case of non-uniform QAM, since a grid value optimized per SNR (orcode-rate of FEC) is used, the sub-constellation separation factor “b”may also use a value optimized per SNR (or code-rate of FEC). That is,capacity of constellation transmitted after MIMO encoding may beanalyzed according to the value “b” and the SNR (or code-rate of FEC) tofind the value “B” for providing maximum capacity at a specific SNR(target SNR).

For example, if NU-16 QAM+NU-16 QAM MIMO and P={1, 3.7}, the new value“a” may be computed by

$a = {{\frac{\sqrt{2}}{2}\left( {3.7 - 1} \right)} + {3.7.}}$

At this time, the value b is set to

$\frac{\sqrt{2}}{2}.$

For example, NU-64 QAM+NU-64 QAM MIMO and P={1, 3.27, 5.93, 10.27}, thenew value “a” may be computed by

$a = {{\frac{\sqrt{2}}{2}\left( {10.27 - 5.93} \right)} + {10.27.}}$

At this time, the value b is set to

$\frac{\sqrt{2}}{2}.$

For example, if NU-256 QAM+NU-256 QAM MIMO and P={1, 1.02528, 3.01031,3.2249, 5.2505, 6.05413, 8.48014, 11.385}, the new value “a” may becomputed by

$a = {{\frac{\sqrt{2}}{2}\left( {11.385 - 8.48014} \right)} + {11.385.}}$

At this time, the value b is set to

$\frac{\sqrt{2}}{2}.$

As described above, the PH-eSM PI and FRFD PH-eSM PI methods may beapplied to symbols mapped to non-uniform QAM. Similarly, the PH-eSMmethod may also be applied to symbols mapped to non-uniform QAM. In thiscase, the value “a” may be determined according to the PH-eSM method. Anequation for determining the value “a” is as follows.

$\begin{matrix}{{a = \frac{{b\left( {P_{m} - P_{m - 1}} \right)} + P_{m} + 1}{{b\left( {P_{m} - P_{m - 1}} \right)} + P_{m} - 1}}{for}{{{2^{n}{QAM}} + {2^{n}{QAM}}},{m = 2^{\frac{n}{2} - 1}}}{for}{2^{n}{QAM}}} & \left\lbrack {{Math}\mspace{14mu} {figure}\mspace{14mu} 31} \right\rbrack\end{matrix}$

This new value “a” may be a value a when the PH-eSM method is applied tosymbols mapped to non-uniform QAM.

b is a sub-constellation separation factor as described above. Asdescribed above, the value “b” may be optimized to suit each SNR (orcode-rate of FEC) by analyzing capacity of the encoded constellation.

For example, if NU-16 QAM+NU-16 QAM MIMO and P={1, 3.7}, the new value“a” may be computed by

$a = {\frac{{\frac{\sqrt{2}}{2}\left( {3.7 - 1} \right)} + 3.7 + 1}{{\frac{\sqrt{2}}{2}\left( {3.7 - 1} \right)} + 3.7 - 1}.}$

At this time, the value b is set to

$\frac{\sqrt{2}}{2}.$

For example, if NU-64 QAM+NU-64 QAM MIMO and P={1, 3.27, 5.93, 10.27},the new value “a” may be computed by

$a = {\frac{{\frac{\sqrt{2}}{2}\left( {10.27 - 5.93} \right)} + 10.27 + 1}{{\frac{\sqrt{2}}{2}\left( {10.27 - 5.93} \right)} + 10.27 - 1}.}$

At this time, the value b is set to

$\frac{\sqrt{2}}{2}.$

For example, if NU-256 QAM+NU-256 QAM MIMO and P={1, 1.02528, 3.01031,3.2249, 5.2505, 6.05413, 8.48014, 11.385}, the new value “a” may becomputed by

$a = {\frac{{\frac{\sqrt{2}}{2}\left( {11.385 - 8.48014} \right)} + 11.385 + 1}{{\frac{\sqrt{2}}{2}\left( {11.385 - 8.48014} \right)} + 11.385 - 1}.}$

At this time, the value b is set to

$\frac{\sqrt{2}}{2}.$

Hereinafter, a method of determining NU-QAN and MIMO encoding parameter“a” in the MIMO encoding method (the PH-eSM PI method and the FRFDPH-eSM PI method) applied to symbols mapped to NU-QAM optimized per SNR(or code-rate of FEC) will be described.

In order to apply the PH-eSM PI method and the FRFD PH-eSM PI method tosymbols mapped to NU-QAM per SNR (or code-rate of FEC), the followingtwo elements should be considered. First, in order to obtain shapinggain, NU-QAM optimized per SNR should be found. Second, the MIMOencoding parameter “a” should be determined in each NU-QAM optimized perSNR.

The MIMO encoding scheme (the PH-eSM PI method and the FRFD PH-eSM PImethod), NU-QAM and MIMO encoding parameter suitable for each SNR may bedetermined through capacity analysis as follows. Here, capacity may meanBICM capacity. The process of determining a NU-QAM and MIMO encodingparameter suitable for each SNR may be performed in consideration ofcorrelated channel and power imbalanced channel.

If computation for capacity analysis at MIMO channel is acceptable, itis possible to determine NU-QAM for optimized MIMO, which providesmaximum capacity at a target SNR.

If computation is not acceptable, NU-QAM for MIMO may be determinedusing NU-QAM optimized for SISO. First, with respect to NU-QAM optimizedfor SISO per SNR (or code-rate of FEC), BER performance comparison maybe performed in a non-power imbalanced MIMO channel environment. ThroughBER performance comparison, NU-QAM for MIMO may be determined fromNU-QAM (FEC code rate 5/15, 6/15, . . . 13/15) optimized for SISO. Forexample, constellation for MIMO at code-rate 5/15 of 12 bpcu(NU-64QAM+NU-64QAM) may be set to NU-64QAM corresponding to SISOcode-rate 5/15. In addition, for example, constellation of MIMO FEC coderate 6/15 may be constellation of SISO FEC code rate 5/15. That is,constellation of SISO FEC code rate 5/15 may suitable for MIMO FEC coderate 6/15.

Once NU-QAM is determined, the MIMO encoding parameter “a” optimized perSNR may be determined at a power imbalanced MIMO channel throughcapacity analysis based on the determined NU-QAM. For example, in the 12bpcu and 5/15 code rate environment, the value a may be 0.1571.

Hereinafter, measurement for performance of MIMO encoding according tothe value a will be described. For performance measurement, BICMcapacity may be measured. Through this operation, the value a capable ofmaximizing BICM capacity is determined.

BICM capacity may be expressed by the following equations.

$\begin{matrix}{{{BICMcap}.} = {\int_{\phi}^{\;}{\left( {\sum\limits_{i}\; {\left( {{\int_{Y}^{\;}{{p\left( {{b_{i} = 0},Y} \right)}\log_{2}\frac{p\left( {{b_{i} = 0},Y} \right)}{{p\left( {b_{i} = 0} \right)}{p(Y)}}{Y}}} + {\int_{Y}^{\;}{{p\left( {{b_{i} = 1},Y} \right)}\log_{2}\frac{p\left( {{b_{i} = 1},Y} \right)}{{p\left( {b_{i} = 1} \right)}{p(Y)}}\ {Y}}}} \right){p(\phi)}}} \right){\phi}}}} & \left\lbrack {{Math}\mspace{14mu} {figure}\mspace{14mu} 32} \right\rbrack \\{\mspace{79mu} \begin{matrix}{{p\left( {{b_{i} = j},Y} \right)} = {{p\left( {\left. Y \middle| b_{i} \right. = j} \right)} \cdot {p\left( {b_{i} = j} \right)}}} \\{= {\sum\limits_{M_{i}}\; {{p\left( {\left. Y \middle| S \right. = M_{j}} \right)} \cdot \frac{1}{M^{2}}}}} \\{= {\sum\limits_{M_{i}}\; {\frac{1}{\pi \; \sigma^{2}}{^{\frac{- {{Y - {H_{PI}{PM}_{j}}}}^{2}}{\sigma^{2}}} \cdot \frac{1}{M^{2}}}}}}\end{matrix}} & \left\lbrack {{Math}\mspace{14mu} {figure}\mspace{14mu} 33} \right\rbrack \\\begin{matrix}{\frac{p\left( {{b_{i} = j},Y} \right)}{{p\left( {b_{i} = j} \right)}{p(Y)}} = \frac{p\left( {\left. Y \middle| b_{i} \right. = j} \right)}{p(Y)}} \\{= \frac{p\left( {\left. Y \middle| b_{i} \right. = j} \right)}{\sum\limits_{j}\; {p\left( {{b_{i} = j},Y} \right)}}} \\{= \frac{\sum\limits_{M_{i}}\; {\frac{1}{\pi \; \sigma^{2}}{^{\frac{- {{Y - {H_{PI}{PM}_{j}}}}^{2}}{\sigma^{2}}} \cdot \frac{2}{M^{2}}}}}{\sum\limits_{j}\; {\sum\limits_{M_{i}}\; {\frac{1}{\pi \; \sigma^{2}}{^{\frac{- {{Y - {H_{PI}{PM}_{j}}}}^{2}}{\sigma^{2}}} \cdot \frac{1}{M^{2}}}}}}}\end{matrix} & \left\lbrack {{Math}\mspace{14mu} {figure}\mspace{14mu} 34} \right\rbrack\end{matrix}$

Here, p(b_(i)=0)=p(b_(i)=1)=0.5. In addition, p(S=Mj)=1/M², p(φ))=1/π.Here, Sε(constellation set) and M may mean a constellation size.

Here, Y may be expressed as follows.

$\begin{matrix}{{\begin{bmatrix}{Y_{1}\left( f_{1} \right)} \\{Y_{2}\left( f_{1} \right)}\end{bmatrix} = {{{\frac{1}{\sqrt{1 + \alpha^{2}}}\begin{bmatrix}1 & {\alpha \cdot ^{j\; \phi}} \\^{j\; \phi} & \alpha\end{bmatrix}}\begin{bmatrix}{X_{1}\left( f_{1} \right)} \\{X_{2}\left( f_{1} \right)}\end{bmatrix}} + \begin{bmatrix}n_{1} \\n_{2}\end{bmatrix}}}\mspace{20mu} {Y = \begin{bmatrix}{Y_{1}\left( f_{1} \right)} \\{Y_{2}\left( f_{1} \right)}\end{bmatrix}}\mspace{20mu} {H_{PI} = {\frac{1}{\sqrt{1 + \alpha^{2}}}\begin{bmatrix}1 & {\alpha \cdot ^{j\; \phi}} \\^{j\; \phi} & \alpha\end{bmatrix}}}\mspace{20mu} {X = \begin{bmatrix}{X_{1}\left( f_{1} \right)} \\{X_{2}\left( f_{1} \right)}\end{bmatrix}}\mspace{20mu} {n = \begin{bmatrix}n_{1} \\n_{2}\end{bmatrix}}} & \left\lbrack {{Math}\mspace{14mu} {figure}\mspace{14mu} 35} \right\rbrack\end{matrix}$

That is, Y=H_(PI)X+n. Here, n may be AWGN. X may be expressed by X=PS asdescribed above. BICM capacity may assume AWGN and individuallyidentically distributed (IID) input. In addition, φ may mean a uniformrandom variable U(0, π). In order to consider a correlated channelenvironment and a power imbalanced channel environment which may occurupon using MIMO, H_(PI) of the above-described equation may be assumed.At this time, an alpha value is a power imbalance (PI) factor and may bePI 9 dB: 0.354817, PI 6 dB: 0.501187 or PI 3 dB: 0.70711 according toPI. Here, Mjε{constellation set] bi=j}.

Through this equation, BICM capacity according to the value a may bemeasured to determine an optimal value a.

That is, the method for determining the MIMO encoding parameter mayinclude two steps as follows.

Step 1. Through BER performance comparison for constellation of SISO FECcode rate, NU-QAM having optimal performance of MIMO FEC code-rate to befound is selected.

Step 2. Based on NU-QAM obtained in Step 1, an encoding parameter “a”having optimal performance may be determined through the above-describedBICM capacity analysis.

The value a according to constellation per code rate is shown in thefollowing table. This is merely an example of the value a according tothe present invention.

TABLE 5 8 bpcu 12 bpcu Code rate Constellation a Constellation a 5/15QAM-16 0 NUQ-64 for CR = 5/15 0.1571 6/15 QAM-16 0.0035 NUQ-64 for CR =5/15 0.1396 7/15 QAM-16 0.1222 NUQ-64 for CR = 6/15 0.2129 8/15 QAM-160.1571 NUQ-64 for CR = 8/15 0.2548 9/15 QAM-16 0.1710 NUQ-64 for CR =11/15 0.2653 10/15  QAM-16 0.1780 NUQ-64 for CR = 12/15 0.2566 11/15 QAM-16 0.1798 NUQ-64 for CR = 12/15 0.2548 12/15  QAM-16 0.1815 NUQ-64for CR = 13/15 0.2583 13/15  QAM-16 0.1815 NUQ-64 for CR = 13/15 0.2583

The PH-eSM PI method can be applied for 8 bpcu and 12 bpcu with 16K and64K FECBLOCK. PH-eSM PI method can use the MIMO encoding parametersdefined in the above table for each combination of a value of bits perchannel use and code rate of an FECBLOCK. Detailed constellationscorresponding to the illustrated MIMO parameter table are describedbelow.

The above table shows constellation and MIMO encoding parameter aoptimized per code rate. For example, in case of 12 bpcu and code rateof 6/15 of MIMO encoding, constellation of NUQ-64 which is used in caseof code rate of 5/15 of SISO encoding may be used. That is, in case of12 bpcu and code rate of 6/15 of MIMO encoding, constellation of coderate of 5/15 of SISO encoding may be an optimal value. At this time, thevalue “a” may be 0.1396.

TABLE 6 10 bpcu Code rate Constellation a 5/15 QAM-16/NUQ-64 for CR =5/15 0 6/15 QAM-16/NUQ-64 for CR = 5/15 0 7/15 QAM-16/NUQ-64 for CR =6/15 0 8/15 QAM-16/NUQ-64 for CR = 8/15 0 9/15 QAM-16/NUQ-64 for CR =11/15 0 10/15  QAM-16/NUQ-64 for CR = 12/15 0 11/15  QAM-16/NUQ-64 forCR = 12/15 0 12/15  QAM-16/NUQ-64 for CR = 13/15 0 13/15  QAM-16/NUQ-64for CR = 13/15 0

For the 10 bpcu MIMO case, PH-eSM PI method can use the MIMO encodingparameters defined in the above table. These parameters are especiallyuseful when there is a power imbalance between horizontal and verticaltransmission (e.g. 6 dB in current U.S. Elliptical pole network). TheQAM-16 can be used for the TX antenna of which the transmission power isdeliberately attenuated. Detailed constellations corresponding to theillustrated MIMO parameter table are described below.

The FRFD PH-eSM PI method can use the MIMO encoding parameters of thePH-eSM PI method defined in the above tables for each combination of avalue of bit per channel use and code rate of an FECBLOCK.

The values “a” of the above table may be determined in consideration ofEuclidean distance and Hamming distance and are optimal in code rate andconstellation. Accordingly, it is possible to obtain excellent BERperformance.

FIG. 108 is a diagram showing MIMO encoding input/output when the PH-eSMPI method is applied to symbols mapped to non-uniform 64 QAM accordingto one embodiment of the present invention.

Even when the FRFD PH-eSM PI according to one embodiment of the presentinvention is applied to symbols mapped to non-uniform QAM, aninput/output diagram similar to this figure may be obtained. If theabove-described new value “a” and the encoding matrix of the MIMOencoding equation are used, the constellation shown in this figure maybe obtained by the MIMO encoder input and output.

In the MIMO encoder output of this figure, sub-constellations may belocated. At this time, a distance between sub-constellations may bedetermined by the above-described sub-constellation separation factor“b”. The MIMO encoded constellations may maintain a non-uniformproperty.

FIG. 109 is a graph for comparison in performance of MIMO encodingschemes according to the embodiment of the present invention.

This graph shows comparison in capacity between MIMO encoding schemes inan 8-bpcu/outdoor environment. The PH-eSM PI and FRFD PH-eSM PI methodsof the present invention exhibit better performance than an existingMIMO encoding scheme (GC, etc.) in terms of capacity. This means thatmore efficient transmission is possible in the same environment ascompared with other MIMO techniques.

FIG. 110 is a graph for comparison in performance of MIMO encodingschemes according to the embodiment of the present invention.

This graph shows comparison in capacity according to MIMO encodingschemes in an 8-bpcu/outdoor/HPI9 environment. The PH-eSM PI and FRFDPH-eSM PI methods of the present invention exhibits better performancethan an existing MIMO encoding scheme (SM, GC, PH-eSM, etc.) in terms ofcapacity. This means that more efficient transmission is possible in thesame environment as compared with other MIMO techniques.

FIG. 111 is a graph for comparison in performance of MIMO encodingschemes according to the embodiment of the present invention.

This graph shows comparison in BER according to MIMO encoding schemes inan 8-bpcu/outdoor/random BI, TI environment. The PH-eSM PI and FRFDPH-eSM PI methods of the present invention exhibits better performancethan an existing MIMO encoding scheme (GC, etc.) in terms of BER. Thismeans that more efficient transmission is possible in the sameenvironment as compared with other MIMO techniques.

FIG. 112 is a graph for comparison in performance of MIMO encodingschemes according to the embodiment of the present invention.

This graph shows comparison in BER according to MIMO encoding schemes inan 8-bpcu/outdoor/HPI9/random BI, TI environment. BER Performance of thePH-eSM PI and FRFD PH-eSM PI methods of the present invention is betterthan that of existing MIMO encoding (SM, GC, PH-eSM, etc.) in terms ofcapacity. This means that more efficient transmission is possible in thesame environment as compared other MIMO techniques.

FIG. 113 is a diagram showing an embodiment of QAM-16 according to thepresent invention.

This figure shows a constellation shape of QAM-16 on a complex plane.This figure shows the constellation shape of QAM-16 for all code rates.

FIG. 114 is a diagram showing an embodiment of NUQ-64 for 5/15 code rateaccording to the present invention.

This figure shows the constellation shape of QAM-64 for 5/15 code rateon a complex plane.

FIG. 115 is a diagram showing an embodiment of NUQ-64 for 6/15 code rateaccording to the present invention.

This figure shows the constellation shape of QAM-64 for 6/15 code rateon a complex plane.

FIG. 116 is a diagram showing an embodiment of NUQ-64 for 7/15 code rateaccording to the present invention.

This figure shows the constellation shape of QAM-64 for 7/15 code rateon a complex plane.

FIG. 117 is a diagram showing an embodiment of NUQ-64 for 8/15 code rateaccording to the present invention.

This figure shows the constellation shape of QAM-64 for 8/15 code rateon a complex plane.

FIG. 118 is a diagram showing an embodiment of NUQ-64 for 9/15 and 10/15code rates according to the present invention.

This figure shows the constellation shape of QAM-64 for 9/15 and 10/15code rates on a complex plane.

FIG. 119 is a diagram showing an embodiment of NUQ-64 for 11/15 coderate according to the present invention.

This figure shows the constellation shape of QAM-64 for 11/15 code rateon a complex plane.

FIG. 120 is a diagram showing an embodiment of NUQ-64 for 12/15 coderate according to the present invention.

This figure shows the constellation shape of QAM-64 for 12/15 code rateon a complex plane.

FIG. 121 is a diagram showing an embodiment of NUQ-64 for 13/15 coderate according to the present invention.

This figure shows the constellation shape of QAM-64 for 13/15 code rateon a complex plane.

FIG. 122 is a view illustrating a null packet deletion block 16000according to another embodiment of the present invention.

An upper part of FIG. 122 is a view illustrating another embodiment ofthe mode adaptation module of the input formatting module describedabove in relation to FIG. 3, and a lower part of FIG. 122 is a viewillustrating specific blocks of the null packet deletion block 16000included in the mode adaptation module.

As described above, the mode adaptation module of the input formattingmodule for processing multiple input streams may independently processthe input streams.

As illustrated in FIG. 122, the mode adaptation module for processingeach of the multiple input streams may include a pre-processing block(splitter), input interface blocks, input stream synchronizer blocks,compensating delay blocks, header compression blocks, null data reuseblocks, null packet deletion blocks, and BB frame header insertionblocks. Operations of the input interface blocks, the input streamsynchronizer blocks, the compensating delay blocks and the BB frameheader insertion blocks are the same as those described above inrelation to FIG. 3 and thus detailed descriptions thereof are omittedhere.

The pre-processing block may split the input TS, IP, GS streams intomultiple service or service component (audio, video, etc.) streams. Inaddition, the header compression block may compress a header of an inputsignal based on a header compression mode. The null packet deletionblock 16000 according to an embodiment of the present invention maydelete input null packets and insert information about the number ofdeleted null packets based on positions thereof, before transmission.Some TS input streams or split TS streams may have a large number ofnull-packets present in order to accommodate VBR (variable bit-rate)services in a CBR TS stream. In this case, in order to avoid unnecessarytransmission overhead, null-packets can be identified and nottransmitted. In the receiver, removed null-packets can be re-inserted inthe exact place where they were originally by reference to a deleted DNPfield that is inserted in the transmission, thus guaranteeing constantbit-rate and avoiding the need for time-stamp (PCR) updating.

As illustrated in the lower part of FIG. 122, the null packet deletionblock 16000 according to an embodiment of the present invention mayinclude a PCR packet check block 16100, a PCR region check block 16200,a null packet detection block 16300 and a null packet spreading block16400. A description is now given of operation of each block.

The PCR packet check block 16100 may determine whether input TS packetsinclude a PCR for synchronizing a decoding timing. In the presentinvention, a TS packet including a PCR may be called a PCR packet.

If the position of a PCR is detected as a result of determination, thePCR packet check block 16100 may change the positions of null packetswithout changing the position of the PCR.

The PCR region check block 16200 may check a TS packet including a PCRpacket and determine whether null packets exist within a range of thesame cycle (i.e., PCR region). In the present invention, a period fordetermining whether a PCR is included may be called a null packetposition reconfigurable region.

The null packet detection block 16300 may check null packets includedbetween input TS packets.

The null packet spreading block 16400 may spread null packets within PCRregion information output from the PCR region check block 16200.

The present invention proposes a method for collecting null packets anda method for distributing null packets as examples of a method forchanging the positions of null packets.

FIG. 123 is a view illustrating a null packet insertion block 17000according to another embodiment of the present invention.

An upper part of FIG. 123 is a view illustrating another embodiment ofthe output processor described above in relation to FIG. 13, and a lowerpart of FIG. 123 is a view illustrating specific blocks of the nullpacket insertion block 17000 included in the output processor.

The output processor illustrated in FIG. 123 may perform a reverseprocedure of the operation performed by the mode adaptation moduledescribed above in relation to FIG. 122.

As illustrated in FIG. 123, the output processor according to anembodiment of the present invention may include BB frame header parserblocks, null packet insertion blocks, null data regenerator blocks,header de-compression blocks, de-jitter buffer blocks, a TS clockregeneration block and a TS recombining block. Operations of the blockscorrespond to reverse procedures of those of the blocks of FIG. 122 andthus detailed descriptions thereof are omitted here.

The null packet insertion block 17000 illustrated in the lower part ofFIG. 123 may perform a reverse procedure of the above-describedoperation performed by the null packet deletion block 16000 of FIG. 122.

As illustrated in FIG. 123, the null packet insertion block 17000 mayinclude a DNP check block 17100, a null packet insertion block 17200 anda null packet generator block 17300.

The DNP check block 17100 may check DNP and acquire information aboutthe number of deleted null packets. The null packet insertion block17200 may receive the information about the number of deleted nullpackets output from the DNP check block 17100 and insert the deletednull packets. In this case, the null packets to be inserted may bepreviously generated by the null packet generator block 17300.

FIG. 124 is a view illustrating a null packet spreading method accordingto an embodiment of the present invention.

FIG. 124(a) illustrates TS packets before the null packet spreadingmethod is used, and FIG. 124(b) illustrates TS packets after the nullpacket spreading method is used.

FIG. 124(c) illustrates Math Figures which express DNP1 and DNP2 basedon the null packet spreading method.

As illustrated in FIG. 124(a), the null packet deletion block 16000according to an embodiment of the present invention may determinewhether input TS packets include a PCR for synchronizing a decodingtiming. That is, if null packet position reconfigurable regioninformation is acquired, a broadcast signal transmission apparatusaccording to an embodiment of the present invention may count a totalnumber of null packets (N_(NP)) included in a corresponding period and atotal number of data packets (N_(TSP)) to be transmitted. As illustratedin FIG. 124(a), the total number of data packets is 8 and the totalnumber of null packets corresponds to 958. AVRnP refers to an averagenumber of null packets spreadable between the data packets within thecorresponding period. As illustrated in FIG. 124(a), AVRnP of thecorresponding period is 119.75.

After that, the null packet deletion block 16000 according to anembodiment of the present invention may spread null packets withinoutput PCR region information. That is, if null packets are deleted, DNPindicating the number of null packets is inserted to a position fromwhich the null packets are deleted. The broadcast signal transmissionapparatus according to an embodiment of the present invention mayperform null packet spreading by calculating DNP1 and DNP2. FIG. 124(b)illustrates null packets spread based on DNP1 and DNP2. DNP1 may becalculated using DNP values inserted to correspond to 1 to NTSP-1 TSpackets and the total number of data packets (N_(TSP)) to betransmitted, based on the Math Figure illustrated in FIG. 124(c). DNP1may have an integer value of the above-described average number of nullpackets.

In addition, DNP2 may be calculated as a remainder not processed byDNP1, based on the Math Figure illustrated in FIG. 124(c). DNP2 may havea value greater than or equal to the value of DNP1 and may be insertedbefore the last TS packet or at the end of the null packet positionreconfigurable region.

The null packet spreading method illustrated in FIG. 124 may be moreeffective to solve the above-described problem in a case when themaximum DNP value for null packets generated due to TS packet splittingexceeds 300.

FIG. 125 is a view illustrating a null packet offset method according toan embodiment of the present invention.

If the number of null packets is excessively large, the number canexceed the maximum DNP value even when the null packet spreading methoddescribed above in relation to FIG. 124 is used.

That is, when an input TS stream is split as illustrated in FIG. 125(a),multiple null packets may be generated. Specifically, in a case whenmultiple TS streams are combined into a big TS stream, when a single TSstream is split based on component levels, or when and a big TS streamis split into video packets and audio packets as in UD service, nullpackets may be periodically inserted. TS input streams or split TSstreams having consecutive TS packets and deleted null packets may bemapped into a payload of BB frame. The BB frame includes a BB frameheader and the payload.

In this case, as described above, if the number of null packets is largeas illustrated in FIG. 125(b), the value of DNP can be equal to orgreater than 290 in some cases.

Accordingly, as illustrated in FIG. 125(c), the null packet deletionblock 16000 according to an embodiment of the present invention maydetermine TS packets to be inserted into the payload of the BB frame anddetermine the most basic DNP value as DNP-offset.

According to an embodiment of the present invention, DNP-offset is theminimum number of DNPs belonging to the same BBF. DNP-offset can betransmitted through the BB frame header. As such, the number of DNPsinserted in front of a TS packet may be reduced to implement efficientTS packet transmission, and a larger number of null packets may bedeleted.

Accordingly, as illustrated in FIG. 125(c), the value of DNP-offset is115, and the first DNP has a value of 0 while the second DNP has a valueof 175 obtained by subtracting 115 from an original value 290. The sameprinciple can also be applied sequentially to the other DNPs.

FIG. 126 is a flowchart illustrating a null packet spreading methodaccording to an embodiment of the present invention.

The null packet deletion block 16000 according to an embodiment of thepresent invention may parse input TS packets for analysis (S20000). Inthis case, the null packet deletion block 16000 according to anembodiment of the present invention may parse the TS packets in units ofthe above-described null packet position reconfigurable region.

After that, the null packet deletion block 16000 according to anembodiment of the present invention may determine whether PCRinformation exists in a corresponding null packet positionreconfigurable region (S20100). In this case, the null packet deletionblock 16000 according to an embodiment of the present invention maydetermine the presence of PCR information by checking a PCR flag of anadaptation field in a header of an input TS packet.

If a PCR value exists as a result of determination, the null packetdeletion block 16000 according to an embodiment of the present inventionmay initialize a counter and related values for null packet spreading(S20200), and count the number of input data TS packets and the numberof null packets (S20300). After that, the null packet deletion block16000 according to an embodiment of the present invention may determinewhether a PCR packet exists (S20400). If a PCR value is not present as aresult of determination, the null packet deletion block 16000 accordingto an embodiment of the present invention may continue to count thenumber of null packets and the number of data TS packets (S20300).

If a PCR value exists as a result of determination, the null packetdeletion block 16000 according to an embodiment of the present inventionmay perform null packet spreading (S20500). In this case, the nullpacket deletion block 16000 according to an embodiment of the presentinvention may calculate the above-described DNP1 and DNP2 values, andmay use the above-described null packet offset method if a correspondingvalue exceeds the maximum DNP value.

FIG. 127 is a conceptual diagram illustrating a protocol stack for thenext generation broadcast system based on hybrid according to anembodiment of the present invention.

The present invention proposes a data link (encapsulation) part shown inFIG. 127, and proposes a method for transmitting MPEG-2 TS (TransportStream) and/or IP (Internet Protocol) packets received from an upperlayer over a physical layer. In addition, the present invention providesa signaling transmission method needed to operate a physical layer. Inaddition, when transmission of a new packet type is considered in anupper layer in the future, the present invention can implement a methodfor transmitting the new packet transmission information to a physicallayer.

The corresponding protocol layer may also be referred to as a data linklayer, an encapsulation layer, a Layer 2, or the like. For convenienceof description and better understanding of the present invention, theprotocol layer will hereinafter be referred to as a link layer. When theterm “protocol layer” is actually applied to the present invention, itshould be noted that the term “protocol layer” may be replaced with theterm ‘link layer’ or may also be called a new name as necessary.

The broadcast system according to the present invention may correspondto a hybrid broadcast signal implemented by combination of an IP(Internet Protocol) centric broadcast network and a broadband network.

The broadcast system according to the present invention may be designedto be compatible with the legacy MPEG-2 based broadcast system.

The broadcast system according to the present invention may correspondto a hybrid broadcast system based on a combination of the IP centricbroadcast network, a broadband network, and/or a mobile communicationnetwork or cellular network.

Referring to FIG. 127, a physical layer may use a physical protocoladopted by a broadcast system such as the ATSC and/or DVB system.

In an encapsulation layer, an IP datagram may be obtained from specificinformation acquired from a physical layer, or the obtained IP datagrammay be converted into a specific frame (e.g., RS frame, GSE-lite, GSE orsignal frame). In this case, the frame may include an aggregate of IPdatagrams.

A fast access channel (FAC) may include specific information (e.g.,mapping information between a service ID and a frame) used for access toa service and/or contents.

A broadcast system according to the present invention may use a varietyof protocols, for example, Internet Protocol (IP), User DatagramProtocol (UDP), Transmission Control Protocol (TCP), ALC/LCT(Asynchronous Layered Coding/Layered Coding Transport), RCP/RTCP (RateControl Protocol/RTP Control Protocol), HTTP (Hypertext TransferProtocol), FLUTE (File Delivery over Unidirectional Transport), etc. Astack between protocols may refer to the structure of FIG. 127.

In the broadcast system of the present invention, data may betransmitted in the form of ISOBMFF (ISO base media file format). ESG(Electrical Service Guide), NRT (Non Real Time), A/V (Audio/Video)and/or general data may be transmitted in the form of ISOBMFF.

Data transmission caused by the broadcast network may include linearcontent transmission and/or non-linear content transmission.

RTP/RTCP based AN, and data (closed caption, emergency alert message,etc.) transmission may correspond to linear content transmission.

RTP payload may be encapsulated and transmitted in the form of an RTP/AVstream including a Network Abstraction Layer (NAL) and/or in the form ofan ISO based media file format. RTP payload transmission may correspondto linear content transmission. If the RTP payload is encapsulated andtransmitted in the form of an ISO based media file format, the RTPpayload may include MPEG DASH media segments for A/V or the like.

FLUTE based ESG transmission, non-timed data transmission, and NRTcontent transmission may correspond to non-linear content transmission.The above-mentioned information may be encapsulated and transmitted inthe form of a MIME type file and/or an ISO based media file format. Ifdata is encapsulated and transmitted in the form of an ISO based mediafile format, this data transmission may conceptually include an MPEGDASH media segment for A/V or the like.

Data transmission over the broadband network may be classified intotransmission of contents and transmission of the signaling data.

Content transmission may include transmission of linear content (AN,data(closed caption, emergency alert messages, etc.), transmission ofnon-linear content (ESG, non-timed data, etc.), and transmission of anMPEG DASH based Media segment (NV, data).

Transmission of the signaling data may include transmission of dataincluding a signaling table (including MPD of MPEG DASH) transmitted onthe broadcast network.

The broadcast system of the present invention may support not onlysynchronization between linear/non-linear contents having beentransmitted over the broadcast network, but also synchronization betweencontent transmitted over the broadcast network and content transmittedover the broadband network. For example, if one UD content is dividedinto the broadcast network and the broadband network and thensimultaneously transmitted over the broadcast and broadband networks,the receiver may coordinate a timeline dependent upon a transmission(Tx) protocol, may synchronize contents of the broadcast network and thebroadband contents, and may reconstruct the synchronized contents intoone piece of UE content.

An application layer of the broadcast system may implement technicalcharacteristics, for example, interactivity, personalization, secondscreen, ACR (automatic content recognition), etc. The above-mentionedtechnical characteristics are of importance to the North Americanbroadcast standard evolved from ATSC 2.0 to ATSC 3.0. For example, HTML5may be used to implement interactivity.

In a presentation layer of the broadcast system of the presentinvention, HTML and/or HTML may be used to identify the space and timerelationship between components or between bidirectional applications.

The broadcast system according to another embodiment may be implementedby addition or modification of the above-mentioned broadcast system, anda detailed description of the individual constituent elements will bereplaced with that of the above-mentioned broadcast system.

The broadcast system according to another embodiment of the presentinvention may include a system structure compatible with the MPEG-2system. For example, the linear/non-linear contents transmitted in thelegacy MPEG-2 system can be received or operated in the ATSC 3.0 system,and the A/V and data processing may be adaptively coordinated accordingto whether data received by the ATSC 3.0 system is an MPEG-2 TS or IPdatagram.

In an encapsulation layer of the broadcast system according to anotherembodiment of the present invention, information/data obtained from aphysical layer may be converted into the MPEG-2 TS or IP datagram, ormay be converted into a specific frame (e.g., RS frame, GSE-lite, GSE orsignal frame, etc.) using the IP datagram.

The broadcast system according to another embodiment may includesignaling information capable of being adaptively obtained according towhether MPEG-2 TS or IP datagram is used to acquire the service/contentthrough the broadcast network. That is, when obtaining signalinginformation from the broadcast system, the signaling information may beobtained on the basis of MPEG-2 TS, or may be obtained from data basedon a UDP protocol.

The broadcast system of the present invention may supportsynchronization between the linear/non-linear contents based on thebroadcast network encapsulated by MPEG-2 TS and/or IP datagram.Alternatively, the broadcast system can support synchronization betweencontent fragments that are respectively transmitted through thebroadcast network and the broadband network. For example, if one UDcontent is divided into the broadcast network and the broadband networkand then simultaneously transmitted over the broadcast and broadbandnetworks, the receiver may coordinate a timeline dependent upon atransmission (Tx) protocol, may synchronize contents of the broadcastnetwork and the broadband contents, and may reconstruct the synchronizedcontents into one piece of UE content.

FIG. 128 is a conceptual diagram illustrating an interface of a linklayer according to an embodiment of the present invention.

Referring to FIG. 128, the transmitter may consider an exemplary case inwhich IP packets and/or MPEG-2 TS packets mainly used in the digitalbroadcasting are used as input signals. The transmitter may also supporta packet structure of a new protocol capable of being used in the nextgeneration broadcast system. The encapsulated data of the link layer andsignaling information may be transmitted to a physical layer. Thetransmitter may process the transmitted data (including signaling data)according to the protocol of a physical layer supported by the broadcastsystem, such that the transmitter may transmit a signal including thecorresponding data.

On the other hand, the receiver may recover data and signalinginformation received from the physical layer into other data capable ofbeing processed in a higher layer. The receiver may read a header of thepacket, and may determine whether a packet received from the physicallayer indicates signaling information (or signaling data) or recognitiondata (or content data).

The signaling information (i.e., signaling data) received from the linklayer of the transmitter may include first signaling information that isreceived from an upper layer and needs to be transmitted to an upperlayer of the receiver; second signaling information that is generatedfrom the link layer and provides information regarding data processingin the link layer of the receiver; and/or third signaling informationthat is generated from the upper layer or the link layer and istransferred to quickly detect specific data (e.g., service, content,and/or signaling data) in a physical layer.

FIG. 129 is a conceptual diagram illustrating a packet structure of alink elayer according to an embodiment of the present invention.

In accordance with an embodiment of the present invention, the packet ofthe link layer may include a fixed header, an extended header, and/orpayload.

A fixed header is designed to have a fixed size. For example, the fixedheader may be 1 byte long. The extended header can be changed in size.Payload including data received from the higher layer may be locatedbehind the fixed header and the extended header.

The fixed header may include a packet type element and/or an indicatorpart element.

The packet type element may be 3 bits long. The packet type element mayidentify a packet type of a higher layer (i.e., a higher layer of thelink layer). The packet type identified by the packet type element valuewill hereinafter be described in detail.

The indicator part element may include information regarding a payloadconstruction method and/or construction information of the extendedheader. The construction method and/or the construction informationindicated by the indicator part element may be changed according topacket types.

FIG. 130 shows packet types dependent upon the packet type elementvalues according to an embodiment of the present invention.

Referring to FIG. 130, if the packet type element is set to ‘000’, thismeans that a packet transferred from the higher layer to the link layeris an IPv4 (Internet Protocol version 4) packet.

If the packet type element value is set to ‘001’, this means that apacket transferred from the higher layer to the link layer is an IPv6(Internet Protocol version 6) packet.

If the packet type element value is set to ‘010’, this means that apacket transferred from the higher layer to the link layer is aCompressed IP packet.

If the packet type element value is set to ‘011’, this means that apacket transferred from the higher layer to the link layer is an MPEG-2TS packet.

If the packet type element value is set to ‘101’, this means that apacket transferred from the higher layer to the link layer is aPacketized Stream packet. For example, the Packetized Stream maycorrespond to an MPEG media transport packet.

If the packet type element value is set to ‘110’, this means that apacket transferred from the higher layer to the link layer is a packetfor transmitting signaling information (signaling data).

If the packet type element value is set to ‘111’, this means that apacket transferred from the higher layer to the link layer is a FramedPacket type.

FIG. 131 is a conceptual diagram illustrating a header structure of alink layer packet when an IP packet is transmitted to the link layeraccording to an embodiment of the present invention.

Referring to FIG. 131, if the IP packet is input to the link layer, thepacket type element value may be 000B (3 bits of 000) or 001B (3 bits of001).

Referring to a packet header of the link layer when an IP packet isinput, the indicator part element located next to the packet typeelement may include a C/S (Concatenation/Segmentation) field and/or anadditional bit of 3 bits (hereinafter referred to as an additionalfield).

In case of the packet of the link layer, an additional field of thefixed header and information of the extended header may be decidedaccording to the CS (Concatenation/Segmentation) field of 2 bits locatedbehind the packet type element.

The C/S field indicates the processing type of the input IP packet, andmay include information regarding the extended header length.

In accordance with an embodiment of the present invention, the case inwhich the C/S field is set to 00B (2 bits of 00) may indicate thatpayload of the link layer packet includes a normal packet. The normalpacket may indicate that the input IP packet is used as payload of thelink layer packet without change. In this case, the additional field ofthe fixed header part is not in use, and may be reserved for asubsequent use. In this case, the extended header may not be used.

If the C/S field is set to ‘01B’ (2 bits of ‘01’), this means thatpayload of the link layer packet includes a concatenated packet. Theconcatenated packet includes one or more IP packets. That is, one ormore IP packets may be contained in payload of the link layer packet. Inthis case, the extended header is not used, and the additional fieldlocated subsequent to the C/S field may be used as the count field. Adetailed description of the count field will hereinafter be described indetail.

If the C/S field is set to ‘10B’ (2 bits of ‘10’), this means thatpayload is composed of segmented packets. The segmented packet isobtained by dividing one IP packet into a few segments. Specifically,the segmented packet may include one segment from among the dividedsegments. That is, payload of the link layer packet may include any oneof a plurality of packets contained in the IP packet. The additionalfield located behind the C/S field is used as the segment ID. Thesegment ID may uniquely identify the segment. The segment ID is assignedwhen the IP packet is segmented. In more detail, if segments to berespectively transmitted in the future are integrated, the segment IDcan indicate the presence of a constituent element of the same IPpacket. The segment ID may be 3 bits long, and at the same time cansupport segmentation of the IP packet. For example, the divided segmentsobtained by one IP packet may have the same segment ID. In this case,the extended header may be 1 byte long. In this case, the extendedheader may include the Seg_SN (Segment Sequence Number) field and/or theSeg_Len_ID (Segment Length ID) field.

The Seg_SN field may be 4 bits long, and may indicate a sequence numberof the corresponding segment for use in the IP packet. When the Seg_SNfield IP packet is segmented, the Seg_SN field may be used to confirmthe order or sequence of each segment. Accordingly, although the linklayer packets including a payload segmented from one IP packet may havethe same segment ID (Seg_ID), the link layer packets may have differentSeg_SN field values. The Seg_SN field may be 4 bits long. In this case,one IP packet can be segmented into a maximum of 16 segments. If a userdesires to divide the IP packet into many more segments, the Seg_SNfield is increased in size so that the Seg_SN field may indicate eachorder of the segment and/or the number of segments.

The Seg_Len_ID (Segment Length ID) field may be 4 bits long, and may beused to identify the segment length. The actual segment length accordingto the Seg_Len_ID field value may be identified by a table to bedescribed later. If the length value of an actual segment is signaledinstead of the Seg_Len_ID field, the Seg_Len_ID field of 4 bits may beextended to the segment length field of 12 bits. In this case, theextended header of 2 bytes may be contained in the link layer packet.

If the C/S field value is set to 11B (2 bits of ‘11’), this means anexemplary case in which payload includes the segmented packet as in thecase in which the C/S field value is set to 10B. However, the C/S fieldof 11B may also indicate that the last segment from among severalsegments divided in one IP packet may be contained in a payload. Whensegments are collected to reconstruct one IP packet, the receiver mayidentify the link layer packet configured to transmit the last segmentusing the C/S field value, and the segment contained in the payload ofthe corresponding packet may be recognized as the last segment. Theadditional field located behind the C/S field may be used as the segmentID. In this case, the extended header may be 2 bytes long. The extendedheader may include the Seg_SN (Segment Sequence Number) field and/or theL_Seg_Len (Last Segment Length) field.

The L_Seg_Len field may indicate the actual length of the last segment.If data is segmented to generate the same-sized data segments in theorder from the front part of the IP packet using the Seg_Len_ID field,the last segment may have a different size as compared to anotherprevious segment. Accordingly, the segment length may be directlyindicated using the L_Seg_Len field. The segment length may be changedaccording to the number of allocated bits of the L_Seg_Len field.However, when allocating the number of bits according to the presentinvention, the L_Seg_Len field may indicate that the last segment is1-4095 bytes long.

That is, if one IP packet is divided into a plurality of segments, theIP packet can be divided into a plurality of segments having apredetermined length. However, the length of, the last segment may bechanged according to the length of the IP packet. Accordingly, thelength of the last segment needs to be signaled independently. Adetailed description of the field having the same name may be replacedwith the above-mentioned description.

FIG. 132 is a conceptual diagram illustrating the meaning and headerstructures according to C/S field values.

Referring to FIG. 132, if the C/S field is set to ‘00’, this means thata normal packet is contained in the payload of the link layer packet andthe additional field is reserved. On the other hand, the extended headermay not be contained in the link layer packet. In this case, a totallength of the header of the link layer packet may be 1 byte.

If the C/S field is set to ‘01’, a concatenated packet is contained inthe payload of the link layer packet and the additional field may beused as the count field. A detailed description of the count field willbe given later. In the meantime, the extended header may not becontained in the link layer packet. In this case, a total length of theheader of the link layer packet may be 1 byte.

If the C/S field is set to ‘10’, the segmented packet may be containedin the payload of the link layer packet, and the additional field may beused as the segment ID. In the meantime, the extended header may becontained in the link layer packet, and the extended header may includethe Seg_SN field and/or the Seg_Len_ID field. A detailed description ofthe Seg_SN field or the Seg_Len_ID field may be replaced with theabove-mentioned description or a description to be given later. A totallength of the link layer packet may be 2 bytes.

If the C/S field is set to ‘11’, the segmented packet (i.e., packetincluding the last segment) may be contained in the payload of the linklayer packet, and the additional field may be used as the segment ID.Meanwhile, the extended header may be contained in the link layerpacket, and the extended header may include the Seg_SN field and/or theL_Seg_Len field. A detailed description of the Seg_SN field or theL_Seg_Len field may be replaced with the above-mentioned description ora description to be described given. A total length of the link layerpacket may be 3 bytes.

FIG. 133 is a conceptual diagram illustrating the meaning according tothe count field values.

Referring to FIG. 133, the count field may be used in the case in whichthe payload of the link layer packet includes a concatenated packet. Thecount field may indicate how many IP packets are contained in onepayload. The value of the count field may indicate the number ofconcatenated IP packets. However, zero or one concatenation has nomeaning, such that the count field may indicate that the IP packets, thenumber of which is denoted by “count field value+2”, are contained inthe payload. In accordance with one embodiment, 3 bits may be allocatedto the count field, so that this means that a maximum of 9 IP packetshas been contained in the payload of the link layer packet. If there isa need to include many more IP packets in one payload, the length of thecount field may be extended, or 9 or more IP packets of the extendedheader may be additionally signaled.

FIG. 134 is a conceptual diagram illustrating the meaning and segmentlengths according to values of Seg_Len_ID field.

Referring to FIG. 134, the Seg_Len_ID field may be used to indicate thelength of segments other than the last segment from among severalsegments. In order to reduce overhead of the header needed forindicating the segment length, an available segment size may be limitedto 16 segments.

The segment length is decided in response to the packet input sizepredetermined by a code rate of Forward Error Correction (FEC) processedby a physical layer, and the decided segment length may be designated asa length for each value of the Seg_Len_ID field. For example, inassociation with each value assigned to the Seg_Len_ID field, thesegment length may be predetermined. In this case, information regardingthe segment length dependent upon each value of the Seg_Len_ID field isgenerated by the transmitter and transmitted to the receiver, such thatthe receiver may store the received information therein. In themeantime, the segment length established to have each value of theSeg_Len_ID field may be changed. In this case, the transmitter maygenerate new information and transmit the new information to thereceiver, and the receiver may update stored information on the basis ofthe above new information.

In the meantime, if the physical layer processing is performedirrespective of the segment length, the segment length may be calculatedas shown in the equation of FIG. 134.

In Equation of FIG. 134, Len_Unit (Length Unit) may be a basic unit forindicating the segment length, and min_Len may be a minimum value of thesegment length. Len_Unit and min_Len may be set to the same value notonly in the transmitter but also in the receiver. After theabove-mentioned parameters of Equation have been decided once, it ispreferable that the above parameters remain unchanged in terms of systemthroughput. This value may be decided in consideration of the FECprocessing throughput of the physical layer during an initiation processof the system. For example, as shown in FIG. 134, the Len_Unit ormin_Len value may indicate the segment length differently represented inresponse to the Seg_Len_ID field value. At this time, the parameterLen_Unit may be 256, and the parameter ‘min_Len’ may be 512.

FIG. 135 is a conceptual diagram illustrating an equation forencapsulating a normal packet and an equation for calculating a linklayer packet length.

Referring to FIG. 135, if the input IP packet is not concatenated orsegmented within the processing range of the physical layer as describedabove, the IP packet may be encapsulated into a normal packet. Thefollowing contents may be equally applied to IPv4 and IPv6 IP packets.One IP packet may be used as payload of the link layer packet withoutchange, the packet type element value may be set to 000B (IPv4) or 001B(IPv6), and the C/S field value may be set to 00B (Normal Packet). Theremaining three bits of the fixed header may be set to a reserved fieldto be used for another usage in future.

The link layer packet length can be identified as follows. A specificfield indicating the IP packet length may be contained in the header ofthe IP packet. The field indicating the length is always located at thesame position, such that the receiver may confirm the field located at aspecific position spaced apart from an initial part (start part) of thelink layer packet by a predetermined offset, such that the payloadlength of the link layer packet can be recognized.

The receiver can read the length field having the length of 2 bytes at aspecific position spaced apart from the start point of the payload by 2bytes in case of IPv4, and can read the length field having the lengthof 2 bytes at a specific position spaced apart from the start point ofthe payload by 4 bytes in case of IPv6.

Referring to FIG. 135, assuming that the IPv4 length field is set toLIPv4, LIPv4 indicates a total length of IPv4. In this case, if theheader length LH (1 byte) of the link layer packet is added to LIPv4,the length of the entire link layer packet is obtained. In this case, LTmay indicate the length of the link layer packet.

Referring to the equation of FIG. 135, assuming that the IPv6 lengthfield is denoted by LIPv6, LIPv6 indicates only the payload length ofthe IPv6 IP packet. Accordingly, if the header length LH (1 byte) of thelink layer packet is added and the fixed header length (40 bytes) ofIPv6 is additionally added, the length of the link layer packet isobtained. Here, LT may denote the length of the link layer packet.

FIG. 136 is a conceptual diagram illustrating a process forencapsulating a concatenated packet and an equation for calculating alink layer packet length.

Referring to FIG. 136, if the input IP packet does not arrive within theprocessing range of the physical layer, some IP packets are concatenatedand encapsulated into one link layer packet. The following descriptioncan also be applied to IP packets of IPv4 and IPv6.

Some IP packets may be used as the payload of the link layer packet, thepacket type element value may be set to 000B (IPv4) or 001B (IPv6), andthe C/S field may be set to 01 B (Concatenated Packet). In addition, thecount field of 3 bits indicating how many IP packets are contained inone payload may be concatenated to the C/S field of 01 B.

In order to calculate the length of the concatenated packet by thereceiver, a similar way to the normal packet case may be used. Assumingthat the number of concatenated IP packets indicated by the count fieldis denoted by n, the header length of the link layer packet is denotedby LH, and the length of each IP packet is denoted by Lk (where 1≦k≦n),the entire link layer packet length (LT) can be calculated as shown inthe equation.

Since the concatenated packet has the fixed header information only,LH=1 (byte) is achieved, and each Lk (where 1≦k≦n) value can beconfirmed by reading the value of the length field contained in theheader of each IP packet contained in the concatenated packet. Thereceiver may parse the length field of a first IP packet at a specificposition that has a predetermined offset on the basis of a payload startposition after the link layer packet header has ended, and may identifythe length of a first IP packet using this length field. The receivermay parse the length field of a second IP packet at a specific positionthat has a predetermined offset on the basis of a length end point ofthe first IP packet, and may identify the length of the second IP packetusing this length field. The above-mentioned operation is repeated apredetermined number of times corresponding to the number of IP packetscontained in the payload of the link layer packet, so that the payloadlength of the link layer packet can be identified.

FIG. 137 is a conceptual diagram illustrating a process for calculatingthe length of a concatenated packet including an IPv4 packet and anequation for calculating an offset value at which a length field of theIP packet is located.

When the IP packet is input to the transmitter, the transmitter has nodifficulty in reading the length field of the IP packet. However, thereceiver can recognize only the number of IP packets constructing thelink layer packet through the header, such that the position of eachlength field is not well known in the art. However, since the lengthfield is always located at the same position of the header of the IPpacket, the position of the length field is detected using the followingmethod, so that the length of each IP packet contained in the payload ofthe concatenated packet can be calculated and recognized.

Assuming that n IP packets contained in the payload of the concatenatedpacket are respectively denoted by IP1, IP2, . . . , IPk, . . . , IPn,the position of the length field corresponding to IPk may be spacedapart from a start point of the payload of the concatenated packet by Pkbytes. In this case, Pk (where 1≦k≦n) may be an offset value at whichthe length field of the k-th IP packet is located on the basis of astart point of the payload of the concatenated packet, and the Pk valuecan be calculated as shown in the equation of FIG. 137.

In this case, P1 of the IPv4 packet is 2 bytes. Therefore, the Pk valueis successively updated from P1, and the Lk value corresponding to thePk value is read. If Lk is applied to the equation of FIG. 136, thelength of concatenated packet can be finally calculated.

FIG. 138 is a conceptual diagram illustrating a process for calculatingthe length of a concatenated packet including an IPv6 packet and anequation for calculating an offset value at which a length field of theIP packet is located.

If the IPv6 packets are concatenated and contained in the payload of thelink layer packet, a method for calculating the payload length is asfollows. The length field contained in the IPv6 packet indicatesinformation regarding the payload length of the IPv6 packet, and 40bytes indicating the length of a fixed header of IPv6 are added to thepayload length of the IPv6 packet indicated by the length field, suchthat the length of IPv6 packet can be calculated.

Assuming that n IP packets contained in the payload of the concatenatedpacket are respectively denoted by IP1, IP2, IPk, . . . , IPn, theposition of the length field corresponding to IPk may be spaced apartfrom the start position of the payload of the concatenated packet by Pkbytes. In this case, Pk (wherein 1 k≦n) may be an offset value at whichthe length field of the k-th IP packet is located on the basis of astart point of the payload of the concatenated packet, and may becalculated by the equation shown in FIG. 138. In this case, P1 in caseof IPv6 has 4 bytes. Accordingly, the Pk value is successively updatedfrom P1, and Lk corresponding to the Pk value is read. If this Lk valueis applied to the equation of FIG. 136, the length of concatenatedpacket can be finally calculated.

FIG. 139 is a conceptual diagram illustrating an encapsulation processof a segmented packet according to an embodiment of the presentinvention.

The following description can be equally be applied to the IPv4 IPpacket and the IPv6 IP packet. One IP packet is segmented to result in apayload of several link layer packets. The packet type element value maybe set to 000B (IPv4) or 001B (IPv6), and the C/S field value may be 10Bor 11B according to the segment construction.

The C/S field may be set to 11B only in a specific segment correspondingto the last part of the IP packet, and may be set to 10B in theremaining segments other than the above specific segment. The C/S fieldvalue may also indicate information of the extended header of the linklayer packet as described above. That is, if the C/S field is set to10B, the header is 2 bytes long. If the C/S field is set to 11B, theheader is 3 bytes long.

In order to indicate the segmentation state from the same IP packet, theSeg_ID (segment ID) values contained in the headers of the individuallink layer packets must have the same value. In order to allow thereceiver to indicate the order (sequence) information of segments forrecombination of normal IP packets, the sequentially increasing Seg_SNvalues are recorded in the header of each link layer packet.

When the IP packet is segmented, the segment length is decided asdescribed above, and the segmentation process based on the same lengthis carried out. Thereafter, the Seg_Len_ID value appropriate for thecorresponding length information is recorded in the header. In thiscase, the length of the last segment may be changed as compared to theprevious segment, so that the length information may be directlydesignated using the L_Seg_Len field.

The length information designated by the Seg_Len_ID field and theL_Seg_Len field may indicate only payload information of the segment(i.e., link layer packet), such that the receiver may identify thelength information of the entire link layer packet by adding the headerlength of the link layer packet to the payload length of the link layerpacket using the C/S field.

FIG. 140 is a conceptual diagram illustrating a segmentation process ofan IP packet and header information of a link layer packet according toan embodiment of the present invention.

When the IP packet is segmented and encapsulated into the link layerpacket, the field values allocated to the header of respective linklayer packets are shown in FIG. 14.

For example, if the IP packet having the length of 5500 bytes in the IPlayer is input to the link layer, this IP packet is divided into 5segments (S1, S2, S3, S4, S5), and headers (H1, H2, H3, H4, H5) areadded to the 5 segments, so that the added results are encapsulated intothe individual link layer packets.

Assuming that the case of using the IPv4 packet is used, the packet typeelement value may be set to 000B. The C/S field value is set to 10B inthe range of H1-H4, and the C/S field value of H5 is set to 11B. All thesegment IDs (Seg_IDs) indicating the same IP packet structure may be setto 000B, and the Seg_SN field is sequentially denoted by 0000B 0100B inthe range of H1-H5.

The resultant value obtained when 5500 bytes is divided by 5 is 1100bytes. Assuming that the segment is composed of the length of 1024 byteslocated closest to the 1100 bytes, the length of the last segment S5 isdenoted by 1404 bytes (010101111100B). In this case, the Seg_Len_IDfield may be set to 0010B as shown in the above-mentioned example.

FIG. 141 is a conceptual diagram illustrating a segmentation process ofan IP packet including a cyclic redundancy check (CRC) according to anembodiment of the present invention.

When the IP packet is segmented and transmitted to the receiver, thetransmitter may attach the CRC to the rear of the IP packet in such amanner that integrity of combined packets can be confirmed by thereceiver, and finally the segmentation process may be carried out.Generally, since CRS is added to the last part of the packet, the CRS iscontained in the last segment after completion of the segmentationprocess.

When the receiver receives data having a length exceeding the length ofthe last segment, the received data may be recognized as CRC.Alternatively, the length including the CRC length may be signaled asthe length of the last segment.

FIG. 142 is a conceptual diagram illustrating a header structure of alink layer packet when MPEG-2 TS (Transport Stream) is input to a linklayer according to an embodiment of the present invention.

The packet type element may identify that the MPEG-2 TS packet is inputto the link layer. For example, the packet type element value may be setto 011 B.

If the MPEG-2 TS is input, the header structure of the link layer packetis shown in FIG. 16. If the MPEG-2 TS packet is input to the link layer,the header of the link layer packet may include the packet type element,the count field, the PI (PID Indicator) field, and/or the DI (DeletedNull Packet Indicator) field.

For example, the 2-bit or 3-bit count field, the 1-bit PI (PIDIndicator) field, and the 1-bit DI (Deleted Null Packet Indicator) fieldmay be arranged subsequent to the packet type of the header of the linklayer packet. If the count field has 2 bits, the remaining 1 bit may beused as a reserved field to be used for a subsequent use in future. Thefixed header part may be constructed in various ways as shown in FIGS.16(a) to 16(d) according to locations of the reserved field. Althoughthe present invention will be disclosed on the basis of the header of(a) for convenience of description and better understanding of thepresent invention, the same description may also be applied to othertypes of headers.

If the MPEG-2 TS packet is input to the link layer (packet type=011),the extended header may not be used.

The count field may indicate how many MPEG-2 TS packets are contained inthe payload of the link layer packet. The size of one MPEG-2 TS packetis greatly less than the size of LDPC (Low-density parity-check) inputindicating the FEC scheme having a high-selection possibility in thephysical layer of the next generation broadcast system, andconcatenation of the link layer can be basically considered. That is,one or more MPEG-2 TS packets may be contained in the payload of thelink layer packet. However, the number of concatenated MPEG-2 TS packetsis limited to some numbers, so that this information may be identifiedby 2 bits or 3 bits. Since the length of the MPEG-2 T packet is fixed toa predetermined size (e.g., 188 bytes), the receiver may also estimatethe payload size of the link layer packet using the count field. Anexample of indicating the number of MPEG-2 TS packets according to thecount field will hereinafter be described in detail.

PI (Common PID indicator) field is set to ‘1’ when the MPEG-2 TS packetscontained in the payload of one link layer packet have the same PIDs(Packet Identifiers). On the contrary, if the MPEG-2 TS packetscontained in the payload of one link layer packet have different PIDs,the PI field is set to ‘0’. The PID field may be 1 bit long.

DI (Null Packet Deletion Indicator) field is set to 1 when a null packetcontained in the MPEG-2 TS packet and then transmitted is deleted. Ifthe null packet is not deleted, the DI field is set to ‘0’. The DI fieldmay be 1 bit long. If the DI field is set to 1, the receiver may reusesome fields of the MPEG-2 TS packet so as to support null packetdeletion in the link layer.

FIG. 143 shows the number of MPEG-2 TS packets contained in a payload ofthe link layer packet according to values of a count field.

If the count field is 2 bits long, the concatenated MPEG-2 TS packetsmay be present in four cases. The payload size of the link layer packetother than synchronous bytes (Sync Bytes) (47H) may also be identifiedby the count field.

The number of MPEG-2 TS packets to be allocated according to the countfield value may be changed according to system designers.

FIG. 144 is a conceptual diagram illustrating a header of the MPEG-2 TSpacket according to an embodiment of the present invention.

Referring to FIG. 144, the header of the MPEG-2 TS packet may include aSync Byte field, a Transport Error Indicator field, a payload unit startindicator field, a transport priority field, a PID field, a transportscrambling control field, an adaptation field control field, and/or acontinuity counter field.

The Sync Byte field may be used for packet synchronization, and may beexcluded in the case of encapsulation at the link layer. A transporterror indicator (EI) located next to the Sync Byte field is not used bythe transmitter, and may be used to inform a higher layer of thepresence of an error incapable of being recovered by the receiver. As aresult, the Transport Error Indicator field is not used by thetransmitter.

The Transport Error Indicator field is established in a demodulationprocess on the condition that it is impossible to correct errors of thestream. In more detail, the Transport Error Indicator field may indicatethe presence of errors incapable of being corrected in the packet.

The payload unit start indicator field may identify whether PES(Packetized elementary stream) or PSI (Program-specific information) isstarted.

The transport priority field may indicate whether the correspondingpacket has a higher priority than other packets having the same PID.

The PID field may identify each packet.

The transport scrambling control field may indicate whether or not ascramble is used, and/or may indicate whether a scramble is used usingan odd or even key.

The adaptation field control field may indicate the presence or absenceof the adaptation field.

The continuity counter field may indicate an order number (or sequencenumber(of the payload packet.

FIG. 145 is a conceptual diagram illustrating a process for allowing atransceiver to change a usage of a transport error indicator fieldaccording to an embodiment of the present invention.

If the DI field is set to 1, the Transport Error Indicator field may beused as a Deletion Point Indicator (DP1) field in the link layer of thetransmitter as shown in FIG. 19. The Deletion Point Indicator (DP1)field may be recovered to the Transport Error Indicator field aftercompletion of the null packet-related processing in the link layer ofthe receiver. That is, the DI field may indicate whether the null packetis deleted, and at the same time may indicate whether the usage of theTransport Error Indicator field of the MPEG-2 TS header is changed.

FIG. 146 is a conceptual diagram illustrating an encapsulation processof the MPEG-2 TS packet according to an embodiment of the presentinvention.

Basically, the MPEG-2 TS packet concatenation is being considered, sothat a plurality of MPEG-2 TS packets may be contained in the payload ofone link layer packet, and the number of MPEG-2 TS packets may bedecided as described above. Assuming that the number of MPEG-2 TSpackets contained in payload of one link layer packet is denoted by N,respective MPEG-2 TS packets may be denoted by Mk (wherein 1≦k≦n).

The MPEG-2 TS packet may include a fixed header of 4 bytes and a payloadof 184 bytes. 1 byte from among the header of 4 bytes is used as theSync Byte, and is always assigned the same value (47H). Accordingly, oneMPEG-2 TS packet ‘Mk’ may include the sync part (S) of 1 byte, a fixedheader part (Hk) of 3 bytes other than the sync byte, and/or the payloadpart (Pk) of 184 bytes (wherein 1≦k≦n).

If the adaptation field is used in the header of the MPEG-2 TS packet,the fixed header part is extended even to the front part of theadaptation field, and the remaining adaptation parts are contained inthe payload part.

Assuming that N MPEG-2 TS packets are denoted by [M1, M2, M3, . . . ,Mn], the N MPEG-2 TS packets are arranged in the form of [S, H1, P1, S,H2, P2, S, Hn, Pn]. The Sync Part is always set to the same value, suchthat the receiver can detect the corresponding position withoutreceiving any signal from the transmitter, and can perform the insertionaction at the detected position. Accordingly, when the payload of thelink layer packet is constructed, the sync part is excluded so that thepacket can be reduced in size. When an aggregate of the MPEG-2 TSpackets having the above arrangement is constructed as the payload ofthe link layer packet, the sync part is excluded, and the header partand the payload part are separated from each other, so that the MPEG-2TS packets are arranged in the form of [H1, H2, . . . , Hn, P1, P2, Pn].

If the PI field value is set to zero ‘0’ and the DI field is set to zero‘0’, the payload length of the link layer packet has ‘(n×3)+(n×184)’bytes. Thereafter, if 1 byte indicating the header length of the linklayer packet is added to the resultant bytes, the entire link layerpacket length can be calculated and obtained. That is, the receiver canidentify the length of the link layer packet through the above-mentionedprocess.

FIG. 147 is a conceptual diagram illustrating an encapsulation processof the MPEG-2 TS packet having the same PID according to an embodimentof the present invention.

If broadcast data is being successively streamed, the MPEG-2 TSscontained in one link layer packet may have the same PDI value. In thiscase, repeated PID values are simultaneously indicated so that the linklayer packet can be reduced in size. In this case, the PI (PIDindicator) field contained in the header of the link layer packet may beused as necessary.

The PI (Common PID Indicator) value of the header of the link layerpacket may be set to ‘1’. As described above, in the case of using NMPEG-2 TS packets [M1, M2, M3, . . . , Mn] within the payload of thelink layer packet, the sync part is excluded, and the header part andthe payload part are separated from each other, so that the MPEG-2 TSpackets may be arranged in the form of [H1, H2, . . . , Hn, P1, P2, . .. , Pn]. In this case, the header parts [H1, H2, . . . , Hn] of theMPEG-2 TS may have the same PID. Although the PID value is indicated andtransmitted only once, the receiver can recover the corresponding datato an original header. Assuming that a common PID is referred to as aCommon PID (CPID) and the header obtained when the PID is excluded fromthe MPEG-2 TS packet header (Hk) is denoted by H′k (where 1≦k≦n), theheader parts [H1, H2, . . . , Hn] of the MPEG-2 TS constructing thepayload of the link layer packet may be reconstructed in the form of[CPID, H′1, H′2, H′n]. This process may be referred to as Common PIDreduction.

FIG. 148 is a conceptual diagram illustrating an equation forcalculating the length of a link layer packet through a Common PIDreduction process and a Common PID reduction process.

Referring to FIG. 148, the header part of the MPEG-2 TS packet mayinclude a PID of 13 bits. If the MPEG-2 TS packets configured toconstruct the payload of the link layer packet have the same PID values,PID is repeated a predetermined number of times corresponding to thenumber of concatenated packets. Accordingly, the PID part is excludedfrom the header parts [H1, H2, . . . , Hn] of the original MPEG-2 TSpacket, so that the MPEG-2 TS packets are reconstructed in the form of[H′1, H′2, H′n], the common PID value is set to the CPID value, and theCIPD may be located at the front of the reconstructed header part.

The PID value has the length of 13 bits, and the stuffing bit may beadded in a manner that the entire packet is configured in the form of abyte unit. The stuffing bits may be located at the front or rear part ofthe CPID. The stuffing bits may be properly arranged according to thestructure of concatenated protocol layer or the system implementation.

In the case of encapsulating the MPEG-2 TS packets having the same PID,the PID is excluded from the header part of the MPEG-2 TS packets andthen encapsulated, and the payload length of the link layer packet canbe calculated as described above.

As shown in FIG. 148, the header of the MPEG-2 TS packet other than theSync Byte is 3 bytes long. If the PID part of 13 bits is excluded,resulting in the implementation of 11 bits. Accordingly, if N packetsare concatenated to implement (n×11) bits, and if the number ofconcatenated packets is set to a multiple of 8, the (n×11) bits have thelength of a byte unit. The stuffing bits having the length of 3 bits areadded to the common PID length of 13 bits, so that the CPID part havingthe length of 2 bytes can be constructed.

Therefore, in the case of using the link layer packet obtained when NMPEG-2 TS packets having the same PID are encapsulated, assuming thatthe length of the header of the link layer packet is denoted by LH, theCPID part has the length of LCPID, and a total length of the link layerpacket is denoted by LT, the LT value can be calculated as shown in theequation of FIG. 148.

In the embodiment of FIG. 21, LH is 1 byte, and LCPID is 2 bytes.

FIG. 149 is a conceptual diagram illustrating the number of concatenatedMPEG-2 TS packets and the length of a link layer packet according tocount field values when Common PID reduction is used.

If the number of concatenated MPEG-2 TS packets is decided, and if allpackets have the same PID, the above-mentioned common PID reductionprocess can be applied, and the receiver can calculate the length of thelink layer packets according to the above-mentioned equation.

FIG. 150 is a conceptual diagram illustrating a process forencapsulating the MPEG-2 TS packet including a null packet according toan embodiment of the present invention.

In order to transmit the MPEG-2 TS packet at a fixed transfer rate, thenull packet may be contained in the transmission (Tx) stream. The nullpacket is used as overhead in terms of a transmission aspect, and thus,although the transmitter does not the null packet, the receiver canrecover this null packet. When the transmitter deletes the null packetand transmits data and the receiver searches for the number of deletednull packets and the location of deleted null packets so as to performdata recovery, the null packet deletion indicator (DI) field located inthe header of the link layer packet may be used. In this case, the DIvalue of the header of the link layer packet may be set to 1.

The encapsulation action when the null packet is located at an arbitraryposition between input Tx streams may be carried in a manner that npackets other than the null packet are sequentially concatenated. Thecount value indicating how many null packets are successively excludedmay be contained in the payload of the link layer packet, and thereceiver may generate the null packet at an original position on thebasis of this count value so that the original position is filled withthe null packet.

Assuming that N MPEG-2 TS packets other than the null packet are denotedby [M1, M2, M3, . . . , Mn], the null packet may appear at any positionbetween the MPEG-2 TS packets (M1˜Mn). The part at which the null packetis counted a predetermined number of times from among 0˜n times mayappear in a single link layer packet. That is, assuming that theappearance number of times of the above part at which the null packet iscounted within one link layer packet is denoted by ‘p’, the range of pis denoted by 0 to n.

If the count value of each null packet is denoted by Cm, the range of mis denoted by 1≦m≦p, and Cm does not exist in case of p=0. Specificinformation indicating where each Cm is located between the MPEG-2 TSpackets may be denoted using a specific field in which the usage of EI(transport error indicator) is changed to DP1 (Deletion Point Indicator)in the header of the MPEG-2 TS packet.

In the present invention, Cm may have the length of 1 byte. If thepacket to be used later has a margin in length, the 1-byte Cm may alsobe extended. Cm of 1 byte may count a maximum of 256 null packets. Theindicator field of the null packets is located at the header of theMPEG-2 TS packet, and the exclusion of a predetermined number of nullpackets corresponding to “(value denoted by Cm)+1” can be calculated.For example, in case of Cm=0, one null packet may be excluded. In caseof Cm=123, 124 null packets are excluded. If the number of contiguousnull packets is higher than 256, the 257-th null packets are processedas normal packets, and the subsequent null packets can be processed assuch null packets according to the above-mentioned method.

As shown in FIG. 24, the null packet is located between the MPEG-2 TSpackets corresponding to Mi and Mi+1. The count value of the MPEG-2 TSpackets is denoted by C1, and the null packet is located between theMPEG-2 TS packets corresponding to Mj and Mj+1. If the count value ofthe MPEG-2 TS packets may be denoted by Cp, the actual transmissionorder may be denoted by [ . . . , Mi, C1, Mi+1, Mj, Cp, Mj+1, . . . ].

When the header part and the payload part of the MPEG-2 TS packet,instead of the null packet, are separated from each other and rearrangedto construct the payload of the link layer packet, the count value Cm(1≦m≦p) of the null packets is located between the header part and thepayload part of the MPEG-2 TS packet. That is, the payload of the linklayer packets are arranged in the form of [H1, H2, Hn, C1, Cp, P1, P2,Pn], and the receiver confirms the count value one byte by one byte inthe order shown in the DP1 field located at Hk, and recovers as manynull packets as the number of confirmed value according to the order oforiginal MPEG-2 TS packets.

FIG. 151 is a conceptual diagram illustrating a step for processing anindicator configured to count a removed null packet and an equation forcalculating the length of a link layer packet in the processing step.

The DP1 field may be established to indicate deletion of the null packetand the presence of a count value associated with the deleted nullpacket. As shown in FIG. 25, if the DP1 field present at Hi from amongthe header of a plurality of MPEG-2 TS packets is set to 1, this meansthat the null packet located between Hi and Hi+1 is excluded andencapsulated, and its associated 1-byte count value is located betweenthe header part and the payload part.

In the above-mentioned process, the length of the link layer packet canbe calculated by the equation shown in FIG. 151. Therefore, in case ofthe link layer packet that has been obtained by encapsulation of nMPEG-2 TS packets through the null packet exclusion process, assumingthat the header length of the link layer packet is denoted by LH, thecount value Cm (1≦m≦p) of the null packets is denoted by LCount, and thetotal length of the link layer packet is denoted by LT, LT can becalculated by the equation of FIG. 151.

FIG. 152 is a conceptual diagram illustrating a process forencapsulating the MPEG-2 TS packet including a null packet according toanother embodiment of the present invention.

In accordance with another embodiment of the encapsulation methodexcluding the null packets, payload of the link layer packet can beconstructed. In accordance with another embodiment of the presentinvention, when the header part and payload part of the MPEG-2 TSpackets are rearranged to construct the link layer packet payload, thecount value Cm (1≦m≦p) of the null packets can be located at the headerpart and the order or sequence of the null packets may remain unchanged.That is, the count value of the null packets may be contained at aspecific point at which individual MPEG-2 TS headers are ended.Accordingly, when the receiver reads a value of the DP1 field containedin each MPEG-2 TS header, the receiver determines completion of thedeletion of null packets, the receiver reads the count value containedat the last part of the corresponding header, and regenerates as manynull packets as the corresponding count value, such that the regeneratednull packets may be contained in the stream.

FIG. 153 is a conceptual diagram illustrating a process forencapsulating the MPEG-2 TS packets including the same packetidentifiers (PIDs) in a stream including a null packet according to anembodiment of the present invention.

The encapsulation process of MPEG-2 TS packets including the same PID(packet identifier) in the stream including the null packet may becarried out by combination of a first process for encapsulating the linklayer packets other than the above null packets and a second process forencapsulating the MPEG-2 TS packets having the same ID into the linklayer packet.

Since an additional PID indicating the null packet is allocated, thecase in which the null packet is contained in the actual transmissionstream is not processed by the same PID. However, after completion ofthe exclusion process of the null packets, only the count value relatedto the null packet is contained in the payload of the link layer packet,the remaining N MPEG-2 TS packets have the same PID, such that the NMPEG-2 TS packets can be processed by the above-mentioned method

FIG. 154 is a conceptual diagram illustrating an equation forcalculating the length of a link layer packet when the MPEG-2 TS packetshaving the same PIDs are encapsulated in a stream including a nullpacket according to an embodiment of the present invention.

In the stream including the null packet, when MPEG-2 TS packets havingthe same PID are encapsulated, the length of the link layer packet canbe calculated through FIG. 148 and/or FIG. 151. The above equations canbe represented by an equation of FIG. 28.

FIG. 155 is a conceptual diagram illustrating a link layer packetstructure for transmitting signaling information according to anembodiment of the present invention.

In order to transmit signaling information before the receiver receivesthe IP packet or the MPEG-2 TS packet in the same manner as in theupdate process of IP header compression information or broadcast channelscan information, the present invention provides packet formats capableof transmitting signaling data (i.e., signaling data) to the link layer.

In accordance with the embodiment of the present invention, if thepacket type element contained in the header of the link layer packet isset to 110B, a section table (or a descriptor) for signaling may becontained in the payload of the link layer packet and then transmitted.The signaling section table may include a signaling table/table sectioncontained in conventional DVB-SI (service information), PSI/PSIP, NRT(Non Real Time), ATSC 2.0, and MH (Mobile/Handheld).

FIG. 156 is a conceptual diagram illustrating a link layer packetstructure for transmitting the framed packet according to an embodimentof the present invention.

Besides the IP packet or the MPEG-2 TS packet, the packet used in ageneral network can be transmitted through the link layer packet. Inthis case, the packet type element of the header of the link layerpacket may be set to 111B, and may indicate that the framed packet iscontained in the payload of the link layer packet.

FIG. 157 shows a syntax of the framed packet according to an embodimentof the present invention.

The syntax of framed packet may include ethernet_type, length, and/orpacket( ).

The ethernet_type which is a 16-bit field shall identify the type ofpacket in the packet( ) field according to the IANA registry. Onlyregistered values shall be used.

The length which is a 16-bit field shall be set to the total length inbytes of the packet( ) structure.

The packet( ) which is variable length field shall contain a networkpacket.

FIG. 158 is a block diagram illustrating a receiver of the nextgeneration broadcast system according to an embodiment of the presentinvention.

Referring to FIG. 158, the receiver according to an embodiment of thepresent invention may include a receiver (not shown), a ChannelSynchronizer 32010, a Channel Equalizer 32020, a Channel Decoder 32030,a Signaling Decoder 32040, a Baseband Operation Controller 32050, aService Map DB 32060, a Transport Packet Interface 32070, a BroadbandPacket Interface 32080, a Common Protocol Stack 32090, a ServiceSignaling Channel Processing Buffer & Parser 32100, an A/V Processor32110, a Service Guide Processor 32120, an Application Processor 32130,and/or a Service Guide DB 32140.

The receiver (not shown) may receive broadcast signals.

The channel synchronizer 32010 may synchronize a symbol frequency withtiming in a manner that signals received at baseband can be decoded. Inthis case, the baseband may indicate a Tx/Rx region of the broadcastsignal.

The channel equalizer 32020 may perform channel equalization of thereceived (Rx) signal. The channel equalizer 32020 may compensate forsignal distortion encountered when the Rx signals are distorted bymultipath, Doppler effect, etc.

The Channel Decoder 32030 may recover the received (Rx) signal into ameaningful transport frame. The channel decoder 32030 may performforward error correction (FEC) of data or transport frame contained inthe Rx signal.

The signaling decoder 32040 may extract and decode signaling datacontained in the received (Rx) signal. Here, the signaling data mayinclude signaling data and/or service information (SI) to be describedlater.

The baseband operation controller 32050 may control baseband signalprocessing.

The Service Map DB 32060 may store signaling data and/or serviceinformation. The service Map DB 32060 may store signaling datacontained/transmitted in the broadcast signal and/or signaling datacontained/transmitted in the broadband packet.

The transport packet interface 32070 may extract the transport packetfrom the transmission (Tx) frame or the broadcast signal. The transportpacket interface 32070 may extract the signaling data or the IP datagramfrom the transport packet.

The broadband packet interface 32080 may receive broadcast-relatedpackets through the Internet. The broadband packet interface 32080 mayextract a packet obtained through the Internet, and combine or extractthe signaling data or A/V data from the corresponding packet.

The common protocol stack 32090 may process the received packetaccording to the protocol contained in the protocol stack. For example,the common protocol stack 32090 may perform processing for eachprotocol, such that it can process the received packet.

The service signaling channel processing buffer & parser 32100 mayextract signaling data contained in the received packet. The servicesignaling channel processing buffer & parser 32100 may scan servicesand/or contents from the IP datagram or the like, and may extractsignaling information related to acquisition of the services and/orcontents, and parse the extracted signaling information. The signalingdata may be located at a predetermined position or channel within thereceived packet. This position or channel may be referred to as aservice signaling channel. For example, the service signaling channelmay have a specific IP address, a UDP Port number, a transmissionsession ID, etc. The receiver may recognize data being transmitted asthe specific IP address, the UDP port number, and the transmissionsession, etc. as signaling data.

The A/V Processor 32110 may perform decoding of the received audio andvideo data, and presentation processing thereof.

The service guide processor 32120 may extract announcement informationfrom the Rx signal, may manage the service guide DB 32140, and providethe service guide.

The application processor 32130 may extract application data containedin the received packet and/or application-associated information, andmay process the extracted data or information.

The service guide DB 32140 may store the service guide data therein.

The above-described steps can be omitted or replaced by steps executingsimilar or identical functions according to design.

Although the description of the present invention is explained withreference to each of the accompanying drawings for clarity, it ispossible to design new embodiment(s) by merging the embodiments shown inthe accompanying drawings with each other. And, if a recording mediumreadable by a computer, in which programs for executing the embodimentsmentioned in the foregoing description are recorded, is designed innecessity of those skilled in the art, it may belong to the scope of theappended claims and their equivalents.

An apparatus and method according to the present invention may benon-limited by the configurations and methods of the embodimentsmentioned in the foregoing description. And, the embodiments mentionedin the foregoing description can be configured in a manner of beingselectively combined with one another entirely or in part to enablevarious modifications.

In addition, a method according to the present invention can beimplemented with processor-readable codes in a processor-readablerecording medium provided to a network device. The processor-readablemedium may include all kinds of recording devices capable of storingdata readable by a processor. The processor-readable medium may includeone of ROM, RAM, CD-ROM, magnetic tapes, floppy discs, optical datastorage devices, and the like for example and also include such acarrier-wave type implementation as a transmission via Internet.Furthermore, as the processor-readable recording medium is distributedto a computer system connected via network, processor-readable codes canbe saved and executed according to a distributive system.

It will be appreciated by those skilled in the art that variousmodifications and variations can be made in the present inventionwithout departing from the spirit or scope of the inventions. Thus, itis intended that the present invention covers the modifications andvariations of this invention provided they come within the scope of theappended claims and their equivalents.

Both apparatus and method inventions are mentioned in this specificationand descriptions of both of the apparatus and method inventions may becomplementarily applicable to each other.

MODE FOR INVENTION

Various embodiments have been described in the best mode for carryingout the invention.

INDUSTRIAL APPLICABILITY

The present invention is available in a series of broadcast signalprovision fields. It will be apparent to those skilled in the art thatvarious modifications and variations can be made in the presentinvention without departing from the spirit or scope of the inventions.Thus, it is intended that the present invention covers the modificationsand variations of this invention provided they come within the scope ofthe appended claims and their equivalents.

1. A method for receiving a broadcast signal comprising: receiving abroadcast signal; parsing a second packet at the received broadcastsignal, wherein the second packet includes a fixed header, an additionalheader having a variable length, and a payload including data for one ormore first packets, wherein the fixed header includes a packet typeelement for identifying a category of the first packet and an indicatorpart for identifying a construction of the payload, wherein theindicator part includes a C/S (Concatenation/Segmentation) field foridentifying whether the payload of the second packet carries a segmentof one first packets a when the one the first packet is divided into twoor more segments or the payload of the second packet carries more thanone first packets that are concatenated; parsing the first packet usingthe fixed header and/or the additional header; and displaying broadcastcontents by processing the parsed first packet.
 2. The method accordingto claim 1, wherein the packet type element identifies that the firstpacket is an IPv4 (Internet Protocol version 4) packet.
 3. The methodaccording to claim 2, wherein the parsing of the second packet at thereceived broadcast signal includes: if the C/S field indicates that ageneral IPv4 packet is contained in the payload of the second packet,parsing a length field contained in a header of the first packet that isspaced apart from a payload start point of the second packet by apredetermined offset; and identifying a total length of the secondpacket by adding a header length of the second packet to a length of thefirst packet indicated by the length field.
 4. The method according toclaim 2, wherein: if the C/S field indicates that a plurality ofconcatenated IPv4 packets are contained in the payload of the secondpacket, the indicator part further includes a count field indicating thenumber of first packets contained in the payload.
 5. The methodaccording to claim 4, wherein the parsing of the second packet at thereceived broadcast signal includes: if the C/S field indicates that aplurality of concatenated IPv4 packets are contained in the payload ofthe second packet, parsing a length field contained in a header of thefirst packet that is spaced apart from a payload start point of thesecond packet by a predetermined offset, and parsing a length fieldcontained in a header of a next first packet that is spaced apart from apayload end point of the first packet identified by the parsed lengthfield by the predetermined offset; repeating the parsing step of thelength field until all the length fields contained in the headers of asmany first packets as the number of first packets identified by thecount field are completely parsed; and identifying a total length of thesecond packet by adding a header length of the second packet to thelengths of the IPv4 packets indicated by the parsed length fields. 6.The method according to claim 1, wherein the packet type elementidentifies that the first packet is an IPv6 (Internet Protocol version6) packet.
 7. The method according to claim 6, wherein the parsing ofthe second packet at the received broadcast signal includes: if the C/Sfield indicates that a general IPv6 packet is contained in the payloadof the second packet, parsing a length field contained in a header ofthe first packet that is spaced apart from a payload start point of thesecond packet by a predetermined offset; and identifying a total lengthof the second packet by adding a sum of a header length of the firstpacket and a header length of the second packet to a length of the firstpacket indicated by the length field.
 8. The method according to claim6, wherein: if the C/S field indicates that a plurality of concatenatedIPv6 packets are contained in the payload of the second packet, theindicator part further includes a count field indicating the number offirst packets contained in the payload.
 9. (canceled)
 10. An apparatusfor receiving a broadcast signal, comprising: a receiver configured toreceive a broadcast signal; a common protocol stack processor configuredto parse a second packet at the received broadcast signal, wherein thesecond packet includes a fixed header, an additional header having avariable length, and a payload including data for one or more firstpackets, wherein the fixed header includes a packet type element foridentifying a category of the first packet and an indicator part foridentifying a construction of the payload, wherein the indicator partincludes a C/S (Concatenation/Segmentation) field for identifyingwhether the payload of the second packet carries a segment of one firstpackets when the one first packet is divided into two or more segmentsor the payload of the second packet carries more than one first packetsthat are concatenated, and configured to parse the first packet usingthe fixed header and/or the additional header; and an A/V (Audio/Video)processor configured to display broadcast contents by processing theparsed first packet.
 11. The apparatus according to claim 10, whereinthe packet type element identifies that the first packet is an IPv4(Internet Protocol version 4) packet.
 12. The apparatus according toclaim 11, wherein the common protocol stack processor, if the C/S fieldindicates that a general IPv4 packet is contained in the payload of thesecond packet, parses a length field contained in a header of the firstpacket that is spaced apart from a payload start point of the secondpacket by a predetermined offset; and identifies a total length of thesecond packet by adding a header length of the second packet to a lengthof the first packet indicated by the length field.
 13. The apparatusaccording to claim 10, wherein: if the C/S field indicates that aplurality of concatenated IPv4 packets are contained in the payload ofthe second packet, the indicator part further includes a count fieldindicating the number of first packets contained in the payload.
 14. Theapparatus according to claim 13, wherein the common protocol stackprocessor, if the C/S field indicates that a plurality of concatenatedIPv4 packets are contained in the payload of the second packet, parses alength field contained in a header of the first packet that is spacedapart from a payload start point of the second packet by a predeterminedoffset, and parses a length field contained in a header of a next firstpacket that is spaced apart from a payload end point of the first packetidentified by the parsed length field by the predetermined offset;repeats the parsing of the length field until all the length fieldscontained in the headers of as many first packets as the number of firstpackets identified by the count field are completely parsed; andidentifies a total length of the second packet by adding a header lengthof the second packet to the lengths of the IPv4 packets indicated by theparsed length fields.
 15. The apparatus according to claim 10, whereinthe packet type element identifies that the first packet is an IPv6(Internet Protocol version 6) packet.
 16. The apparatus according toclaim 15, wherein the common protocol stack processor, if the C/S fieldindicates that a general IPv6 packet is contained in the payload of thesecond packet, parses a length field contained in a header of the firstpacket that is spaced apart from a payload start point of the secondpacket by a predetermined offset; and identifies a total length of thesecond packet by adding a sum of a header length of the first packet anda header length of the second packet to a length of the first packetindicated by the length field.
 17. (canceled)
 18. (canceled)